Auto-configured equalizer

ABSTRACT

An signal communication device having an auto-configured equalizer. The signal communication device includes a scoping circuit, buffer circuit, select circuit and equalizing circuit. A test signal is transmitted to the signal communication device via a signal path. The scoping circuit generates a waveform trace of the test signal, the waveform trace indicating time interval between receipt of a transition in the test signal and receipt of a reflection of the transition. The buffer circuit stores a plurality of data values that correspond to data signals received via the signal path. The select circuit selects one of the plurality of data values from the buffer circuit based on the time interval indicated by the waveform trace. The equalizing circuit generates an equalizing signal based on the selected data value.

FIELD OF THE INVENTION

The present invention relates generally to high speed signaling withinand between integrated circuit devices, and more particularly toreducing latent signal distortions in high speed signaling systems.

BACKGROUND

Equalizing driver circuits are often used in high speed signalingsystems to mitigate the effects of inter-symbol interference andcrosstalk. Referring to signaling system 100 of FIG. 1, for example,data values queued in buffer 104 are output to signal path 102 by outputdriver 101 simultaneously with transmission of an equalizing signal byequalizing driver 109. In the example shown, the equalizing driver 109includes a shift register 113 and a bank of output drivers 111 togenerate an equalizing signal based on the two most recently transmitteddata values and the data value to be transmitted after the present,reference value. Thus, the equalizing driver 109 constitutes a three-tap(i.e., three data source) equalizer for reducing inter-symbolinterference that results from dispersion of signals transmitted near intime to the reference value (i.e., dispersion-type ISI).

While the equalizing driver 109 is effective for reducing relativelylow-latency distortions such as dispersion-type ISI, other types ofsystematic distortions, such as signal reflections (also referred to asreflection-type ISI), tend to have a much higher latency (i.e., occurmuch later in time relative to transmission of the reference value) andtherefore would require a substantially larger number of taps and acorrespondingly larger shift register to counteract. For example, in thesystem of FIG. 1, a first reflection, A_(T), occurs when a referencesignal encounters an impedance discontinuity at a transmit-sideinterface 105 between a transmit-side portion (102A) and a backplaneportion (102B) of the signal path 102 (e.g., a connector interface to abackplane). Because the reflection bounces between the interface 105 andthe output of the transmit circuit, the reflection will arrive at theinput of a receiver 103 with a latency (i.e., delay relative to arrivalof the unreflected reference signal) equal to approximately twice thereflection flight time between the transmit-side interface 105 and thetransmit circuit output. Impedance discontinuities at the input toreceiver 103 and at a receive-side interface 107 between a receive-sideportion (102C) and the backplane portion (102B) of the signal path 102similarly produce reflections, A_(R), C_(T), C_(R) and D that arrive atthe receiver 103 at respective, latent times according to the additionaldistance traveled by the reflections. FIG. 2 is a waveform diagram ofreflections A_(T), A_(R), B, C_(T), C_(R) and D illustrating theirrespective latencies relative to reference signal arrival time, T (A2_(TR) corresponds to additional reflections produced by the interface105). Because such reflections may occur at latencies on the order oftens or even hundreds of signal transmission intervals, the shiftregister 113 would need to be substantially deeper in order to store thetap values needed to mitigate the resulting distortions. Moreover, theprecise time at which reflections arrive at the receiver 103 aredependent upon system configuration, meaning that a generally applicableequalizer, whether implemented on the transmit or receive side of thesignaling system 100, would need a relatively large number of equalizingtaps to be able to compensate for a reflection occurring at any timebetween the signal transmit time and a worst case latency.Unfortunately, each additional equalizing tap increases the parasiticcapacitance of the transmit or receive circuit, degrading the frequencyresponse of the circuit and potentially increasing the impedancediscontinuity (and therefore the magnitude of reflected signal) at thecircuit input/output.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is illustrated by way of example, and not by wayof limitation, in the figures of the accompanying drawings and in whichlike reference numerals refer to similar elements and in which:

FIG. 1 illustrates a prior-art signaling system;

FIG. 2 is a waveform diagram of reflected signals produced by theprior-art signaling system of FIG. 1;

FIG. 3 illustrates a signaling system according an embodiment of theinvention;

FIG. 4 illustrates an exemplary relationship between clock and datasignals in the signaling system of FIG. 3;

FIG. 5 illustrates the manner in which pre-emphasis and selectable-tapequalization are employed to reduce low- and high-latency distortions inthe signaling system of FIG. 3;

FIG. 6 illustrates a transmit device having circuitry for selectingbetween temporal equalization and cross-talk cancellation data sources;

FIG. 7 illustrates transmit and receive devices configured to performnear-end cross-talk cancellation;

FIG. 8 illustrates a transceiver device that includes both an equalizingtransmitter and an equalizing receiver;

FIG. 9 illustrates an equalizing transceiver according to an embodimentin which both transmitted and received data values are stored andselectively used to source equalizer taps;

FIG. 10 illustrates an exemplary buffer that may be used within thereceiver of FIG. 3;

FIG. 11 is a flow diagram of an exemplary method of selecting a datavalue having desired symbol latency from the buffer of FIG. 10;

FIG. 12 illustrates an exemplary embodiment of a tap select circuit;

FIG. 13 illustrates an exemplary embodiment of the select logic of FIG.12;

FIG. 14 illustrates a generalized select circuit that may be used toselect Q tap values from the buffer circuit of FIG. 12;

FIG. 15 illustrates an embodiment of a switch element that may be usedwithin the switch matrix of FIG. 14;

FIG. 16 illustrates an embodiment of an equalizing receiver;

FIG. 17 illustrates the receive circuit of FIG. 16 in greater detail;

FIG. 18 illustrates an exemplary timing relationship between clock, dataand equalization signals in the equalizing receiver of FIG. 16;

FIG. 19 illustrates a current-sinking output driver that may be usedwithin the equalizing receiver of FIG. 16;

FIG. 20 illustrates an embodiment of a push-pull type of sub-drivercircuit that may be used within an equalizing output driver;

FIG. 21 illustrates another embodiment of a sub-driver circuit that maybe used within an equalizing output driver;

FIG. 22 illustrates an alternative type of equalizing circuit that maybe used in embodiments of the invention;

FIG. 23 illustrates an embodiment of a level shifting circuit that maybe used within the equalizing circuit of FIG. 22;

FIG. 24 illustrates another type of equalizing circuit that may be usedin embodiments of the invention;

FIG. 25 illustrates an embodiment of a level shifting circuit that usedwithin the equalizing circuit of FIG. 24;

FIG. 26 illustrates an equalizing receiver according to an embodiment ofthe invention;

FIG. 27 illustrates a shift register and tap selector that may be usedwithin the equalizing receiver of FIG. 26;

FIG. 28 illustrates an equalizing receiver for receiving a double datarate, multilevel input signal according to an embodiment of theinvention;

FIG. 29 illustrates an exemplary encoding of bits according to the levelof a sampled, multilevel input signal;

FIG. 30 illustrates an exemplary timing relationship between clock, dataand equalization signals in an equalizing receiver;

FIG. 31 illustrates an embodiment of an equalizing receiver thatgenerates receive and equalization clock signals having the phaserelationship shown in FIG. 30;

FIG. 32 illustrates the use of embedded scoping to generate a trace of adata signal over a single symbol time;

FIG. 33 illustrates an embodiment of a signaling system that employsembedded scoping to determine equalizer tap selections, tap weights andtap polarities;

FIG. 34 illustrates an exemplary trace record for a pulse waveformcaptured by an embedded scope within the signaling system of FIG. 33;

FIG. 35 illustrates a method of setting equalization coefficients in asignaling system according to the invention; and

FIG. 36 illustrates a signaling system that employs path length symmetryto reduce the total number of equalization taps needed to compensate forreflection-type ISI.

DETAILED DESCRIPTION

In the following description, specific nomenclature is set forth toprovide a thorough understanding of the present invention. However, itwill be apparent to one skilled in the art that these specific detailsmay not be required to practice the present invention. In someinstances, the interconnection between circuit elements or circuitblocks may be shown as multi-conductor or single conductor signal lines.Each of the multi-conductor signal lines may alternatively be singlesignal conductor lines, and each of the single conductor signal linesmay alternatively be multi-conductor signal lines. A signal is said tobe “asserted” when the signal is driven to a low or high logic state (orcharged to a high logic state or discharged to a low logic state) toindicate a particular condition. Conversely, a signal is said to be“deasserted” to indicate that the signal is driven (or charged ordischarged) to a state other than the asserted state (including a highor low logic state, or the floating state that may occur when the signaldriving circuit is transitioned to a high impedance condition, such asan open drain or open collector condition). A signal driving circuit issaid to “output” a signal to a signal receiving circuit when the signaldriving circuit asserts (or deasserts, if explicitly stated or indicatedby context) the signal on a signal line coupled between the signaldriving and signal receiving circuits. A signal line is said to be“activated” when a signal is asserted on the signal line, and“deactivated” when the signal is deasserted. Additionally, the prefixsymbol “/” attached to signal names indicates that the signal is anactive low signal (i.e., the asserted state is a logic low state). Aline over a signal name (e.g., ‘ <signal name>’) is also used toindicate an active low signal. Active low signals may be changed toactive high signals and vice-versa as is generally known in the art.

Signaling systems and circuits for equalizing low- and high-latencysignal distortions are disclosed herein in various embodiments (herein,equalizing refers to counteracting, canceling or otherwise reducingsignal distortion). In one embodiment, low-latency distortions (e.g.,dispersion-type ISI, cross-talk, etc.) are reduced by a transmit-sideequalization circuit, and high-latency distortions (e.g., signalreflections) are reduced by a receive-side equalization circuit; thelatency of receive-side equalization taps being offset relative to thereception time of a reference signal by the number of transmit-sideequalization taps.

Because data values received within an equalizing receiver are storedfor parallel transfer to application logic, stored data is available tosupply receive-side equalizer taps; no additional shift register orother storage circuit is necessary to store equalizer data. In oneembodiment, a select circuit is provided to selectively route arelatively small subset of the stored data values to equalizing tapswithin the equalizing receiver. By this arrangement, reflected signalsarriving at various, latent times may be counteracted by routing ofselected stored data values to the receive-side equalization taps.Because the number of equalizing taps within the equalizing receiver issmall relative to the range of time for which distortion events aremitigated, the parasitic capacitance of the equalizing receiver is smallrelative to the parasitic capacitance that would result from providing adedicated tap for each stored data value.

Signaling System with Selectable-Tap Equalizer

FIG. 3 illustrates a signaling system 117 according to an embodiment ofthe invention. The system 117 includes an equalizing transmitter 115 andequalizing receiver 116 coupled to one another via a high-speed signalpath 122, and a controller 141 coupled to the transmitter 115 and thereceiver 116 via relatively low-speed signal paths 142A and 142B,respectively. In one embodiment, the signal path 122 is formed bycomponent signal paths 122A, 122B and 122C (e.g., transmission linesthat introduce respective, nonzero propagation delays and exhibitrespective impedance characteristics), each disposed on respectivecircuit boards that are coupled to one another via circuit boardinterfaces 125 and 127 (e.g., connectors). In a specific implementation,signal path 122B is formed on a backplane and signal paths 122A and 122Care formed on respective daughterboards (e.g., line cards) that areremovably coupled to the backplane via connectors 125 and 127. Thetransmitter 115 and receiver 116 are implemented in respectiveintegrated circuit (IC) devices that are mounted on the daughterboards.The controller, which may be a general or special purpose processor,state machine or other logic circuit, is implemented within a thirdintegrated circuit device mounted to a yet another circuit board. In theembodiment of FIG. 3, signal paths 142A and 142B are used to conveyconfiguration information from the controller 141 to the transmitter 115and receiver 116, respectively, and may be disposed on the same circuitboard (or circuit boards) as signal path 122 or implemented by analternative structure such as a cable. The controller may alternativelybe coupled to the transmitter 115 and receiver 116 by a shared signalpath such as a multi-drop bus. The operation of the controller 141 isdiscussed in greater detail below. In alternative embodiments, the ICdevices containing the transmitter 115, receiver 116 and controller 141may be mounted to a common structure with the signal paths 122, 142A and142B coupled directly to the IC devices (e.g., all three ICs mounted toa circuit board and coupled to one another via circuit board traces, orall three ICs packaged within a single multi-chip module with signalpaths 122 and 142 formed between the ICs by bond wires or otherconducting structures). Also, the transmitter 115, receiver 116 andcontroller 141, or any subset thereof, may be included within the sameIC device (e.g., system on chip) and the signal paths 122 and/or 142implemented by a metal layer or other conducting structure within the ICdevice.

The transmitter 115 transmits data on the signal path 122 duringsuccessive time intervals, referred to herein as symbol times. In oneembodiment, illustrated by the timing diagram of FIG. 4, each symboltime, T_(S), corresponds to a half cycle of a transmit clock signal,TCLK, such that two data values (e.g., values A and B) are transmittedon signal path 122 per transmit clock cycle. The transmitted data signalarrives at the input of receiver 116 after propagation time, T_(P), andis sampled by the receiver 116 in response to edges of a receive clocksignal, RCLK. Still referring to FIG. 4, the receive clock signal has aquadrature phase relation to data valid windows (i.e., data eyes) in theincoming data signal such that each sample is captured at the midpointof a data eye. In alternative embodiments, the sampling instant may beskewed relative to data eye midpoints as necessary to satisfy signalsetup and hold time requirements in the receiver 116. Also, more orfewer symbols may be transmitted per cycle of the transmit clock signal.

The equalizing transmitter 115 includes a transmit shift register 124,output driver 121 and equalizer circuit 129; the equalizer circuit 129itself including a shift register 120 and a bank of output drivers 131.At the start of each symbol time, the data value at the head of thetransmit shift register 124, referred to herein as the primary datavalue, is driven onto the signal path 122 by the output driver 121, andthe equalizer circuit 129 simultaneously drives an equalizing signalonto the signal path 122. This type of equalization is referred toherein as transmit preemphasis. In one embodiment, the signal drivenonto the signal path 122 by the output driver 121 (referred to herein asthe primary signal) is a multi-level signal having one of four possiblestates (e.g., defined by four distinct signal ranges) and thereforeconstitutes a symbol representative of two binary bits of information.In alternative embodiments, the primary signal may have more or fewerpossible states and therefore represent more or fewer than two binarybits. Also, the primary signal may be single-ended or differential (anadditional signal line is provided to carry the complement signal in thedifferential case), and may be a voltage or current mode signal.

Each of the output drivers 131 within the equalizer circuit 129 formeither a pre-tap driver or post-tap driver according to whether thesource data value has already been transmitted (post-tap data) or is yetto be transmitted (pre-tap data). In the specific embodiment of FIG. 3,the equalizer includes N post-tap drivers sourced by data values withinthe shift register 120 and one pre-tap driver sourced by a data valuewithin the transmit shift register 124. Accordingly, the resultantequalizing signal driven onto the signal path 122 will have a signallevel according to data values having symbol latencies of −1, 1, 2, . .. , N, where the symbol latency of a given data value refers to thenumber of symbol times by which transmission of the data value precedesthe transmission of the primary value. Different numbers of post-tap andpre-tap drivers may be provided in alternative embodiments, therebyallowing for equalization based on values having different symbollatencies.

Still referring to FIG. 3, the receiver 116 includes a sampling circuit123, buffer 132, tap select circuit 128 and tap select logic 139. Datasignals are sampled by the sampling circuit 123, then stored in thebuffer 132 for eventual use by application logic (not shown). Becausethe buffered data is stored for at least a predetermined time, andrepresents historical data up to a predetermined number of symbollatencies, the buffered data forms an ideal source of post-tap datavalues. That is, in contrast to transmit-side buffering of post-tap datavalues (which requires dedicated storage such as shift register 120),buffering of received data in receiver 116 incurs no additional storageoverhead because the received data values are buffered in any event tofacilitate transfer to receive-side application logic. Additionally, thetap select circuit 128 enables a subset of data values within thebuffered data to be selected to source equalizer taps in a receive-sideequalizer circuit. Because the subset of data values may be selectedaccording to the precise symbol latencies of reflections and otherhigh-latency distortions, a relatively small number of data values maybe selected to form receive-side equalization taps having latencies thatmatch the latencies of the distortions. By this arrangement, highlatency distortions may be reduced by receive-side equalization withoutdramatically increasing the parasitic capacitance of the receiver (i.e.,as would result from a large number of receive-side equalization taps).

In one embodiment, the tap select logic is a configuration circuit thatoutputs a tap select signal according to a configuration value. Asdiscussed below, the configuration value may be automatically generatedby system 117 (e.g., at system startup) or may be empirically determinedand stored within the configuration circuit or elsewhere within system117.

Still referring to FIG. 3, numerous alternative types of equalizationcircuits may be used within the receiver 116. For example, in oneembodiment, the receiver 116 includes an output driver 140 (illustratedin dashed outline in FIG. 3 to indicate its optional nature) to drive anequalizing signal onto the signal path 122 (and therefore to the inputof the sampling circuit 123) coincidentally with the symbol time of anincoming signal. In another embodiment, the sampling circuit 123includes a preamplifier having an equalizing subcircuit. In yet anotherembodiment, an equalizing subcircuit is coupled to the sampling circuititself. Each of these embodiments is described in further detail below.

Still referring to FIG. 3, the distribution of low- and high-latencyequalization functions between the equalizing transmitter 115 andequalizing receiver 116 is achieved through use of a dead range withinthe receive-side buffer 132. That is, the range of stored data valuesthat may be selected to source receive-side equalization taps (i.e., R)is offset from the sampling instant by a number of symbol times, M. Inone embodiment, M is equal to N, the number of post-tap drivers, suchthat transmit preemphasis is used to reduce distortions resulting fromsymbol transmissions up to N symbol times prior to transmission of theprimary signal, and receive-side equalization is used to reducedistortions resulting from symbol transmissions more than N symbol timesprior to transmission of the primary signal. For example, if there arefour post-tap drivers in the transmitter 116 (i.e., M=N=4), then thelowest latency value within the range, R, of stored data values is M+1=5symbol times, and the receiver 116 is said to have a dead range of foursymbol times. In the embodiment of FIG. 3, buffer 132 is formed by ashift register having a dead range component 133 and a selectable-rangecomponent 135, the tap selector 128 being coupled to theselectable-range component 135 to select the subset of tap data sourcestherefrom. In alternative embodiments, the dead range component of thebuffer 132 may include fewer than M storage elements or even zerostorage elements, depending on the time required to receive data andtransfer data into the buffer 132. Also, the tap selector 128 may becoupled to one or more storage elements within the dead range component133 to enable the size of the dead range to be programmed according tothe configuration of the transmit circuit 115. Finally, as discussedbelow, the buffer 132 may include one or more parallel registers inaddition to (or instead of) the shift register formed by components 133and 135.

FIG. 5 illustrates the manner in which pre-emphasis at the transmitter115 and selectable-tap equalization within the receiver 116 are employedto reduce low- and high-latency distortions in the signaling system ofFIG. 3. Waveform 148 depicts the state of the signal path 122 during andafter non-equalized transmission of a primary signal to illustrate thelow- and high-latency distortions that may result. The primary signal istransmitted during a transmit interval 149 (i.e., a symbol time) thatstarts at time T, and the corresponding primary value is used togenerate a transmit-side equalization signal (i.e., preemphasis signal)over a window of N symbol times following the transmit interval 149. Thetransmit-side equalization signal is used to reduce low-latencydistortions that may result from any number of sources including,without limitation, dispersion-type ISI, inductive and capacitivecoupling (which may be compensated, for example, by sourcing apre-emphasis output driver within bank 131 with a value beingtransmitted on a neighboring signal path), and low-latency reflections(e.g., reflections that do not travel significantly further than theunreflected primary signal and therefore arrive at the receiver shortlyafter the primary signal). The primary signal is sampled by the receiver116 during a reception interval (i.e., data valid window) thatcorresponds to the transmit interval 149, the reception interval beingshifted relative to the transmit interval according to the signal flighttime between the transmitter 115 and receiver 116. The selectable-tapequalizer within the receiver 116 has a dead range of M symbol times anda selectable range of R symbol times. Accordingly, the sampled primaryvalue (i.e., the primary received during the reception interval) isselectable to source an equalizer tap within the receiver 116 when thesymbol latency of the sampled primary value is greater than M symboltimes and less or equal to R symbol times. Thus, during given receptioninterval, previously received values having symbol latencies rangingfrom M+1 to R may be selected by the tap selector 128 of FIG. 3 and usedto reduce high-latency distortions. Intervals 150 ₁, 150 ₂, and 150 ₃within interval 153 illustrate equalization windows achieved by tapselections within the tap selector 128. For example, interval 150 ₁corresponds to one or more tap selections used to equalize a distortionoccurring shortly after the dead range, while interval 150 ₃ correspondsto one or more tap selections used to reduce a distortion caused by asignal transmission dozens or even hundreds of symbol times prior to thecurrent reception interval. In the transmit circuit 115 of FIG. 3 andother equalizing transmitters disclosed herein, the polarity of signalcontributions which form the transmit preemphasis signal (including anycross-talk cancellation component thereof) may be fixed or programmableand may be established (or controlled) within the data shift registers(i.e., 124 and 120) or by the output drivers themselves (e.g., outputdrivers within bank 131). Similarly, in the receive circuit 116 of FIG.3 and other equalizing receivers disclosed herein, the polarity ofsignal contributions which form the receiver equalization signal(including any cross-talk cancellation component thereof) may be fixedor programmable and may be established (or controlled) within a datastorage circuit (i.e., buffer 132) or within a receiver equalizationcircuit.

The ability to control tap data latencies with the tap select logic 139and tap selector 128 of FIG. 3 enables the equalization windows 150 tobe shifted within the selectable range, R, as necessary to reducehigh-latency distortions, thereby permitting generalized application ofsystem 117 in environments having a variety of different distortioncharacteristics. In the signaling system 117 of FIG. 3, the controller141 is used to configure one or more of the values of N, M and R (i.e.,the number of transmit-side post-tap equalizers, the receive-side deadrange and the receive-side selectable range) according to system needs.In one embodiment, the controller includes a non-volatile memory tostore empirically or analytically determined values of N, M and R.Alternatively, the signaling system 117 may include a separate storage(e.g., flash memory, or other non-volatile media) to store values of N,M and R (or values that may be used to determine N, M and R), thecontroller 141 being coupled to access such separate storage via signalpath 142 or another path. In either case, when the signaling system 117is initially powered on, the controller 141 communicates the post-tapequalizer count, N, to the transmitter 115 and the dead range andselectable range values, M and R, to the receiver 116. Alternatively,the values of N, M and R may be determined at production time (e.g.,through system testing) or design time, and pre-programmed intoconfiguration circuitry within the transmitter 115 and/or receiver 116,or fixed by design of the transmitter 115 and/or receiver 116. In suchembodiments, the controller 141 and signal path 142 may be omittedaltogether.

As discussed below, embodiments of the invention may additionallyinclude circuitry to automatically determine distortion latencies and toselect correspondingly latent data tap sources from the buffer 132, thusproviding a system-independent solution for reducing systematicdistortion events of virtually any latency. The controller 141 may beused to coordinate operation of the transmitter 115 and receiver 116during such automatic distortion latency determination, and also todetermine appropriate settings of N, M and R based on such distortionlatencies.

Far-End and Near-End Cross-Talk Cancellation

As discussed above in reference to FIG. 3, the transmit-side equalizercircuit 129 may be used to reduce signal distortion resulting frominductive and capacitive coupling of signals transmitted on neighboringsignal paths; a type of equalization referred to as far-end cross-talkcancellation. In one embodiment, the output driver bank 131 includesadditional output drivers to generate equalization signals based onvalues being transmitted on signal paths that are adjacent or otherwiseproximal to the signal path 122. By appropriate polarity control(performed, for example, within the output drivers or data shiftregister), an equalizing signal having a polarity opposite that of aninterfering neighboring signal is transmitted on the signal path 122,thereby reducing the signal interference.

The number of equalizer taps needed for cross-talk cancellation within agiven signaling system is dependent on the physical layout of signalpaths relative to one another. For example, in a system in which signalpaths 122 are arranged relative to one another such that cross-talkinterference is negligible (e.g., paths 122 are spaced apart, arrangedin an orthogonal disposition (e.g., twisted pair), etc.), no equalizertaps may be needed for cross-talk cancellation. By contrast, in a systemin which signal paths form parallel adjacent paths (e.g., paralleltraces on a printed circuit board or parallel conductors within amulti-conductor cable), one or more equalizer taps may be needed foreach adjacent pair of signal paths. In one embodiment of the invention,equalizer taps are selectively coupled to either pre-tap, post-tap orcross-talk cancellation data sources (i.e., primary value beingtransmitted on neighboring path). By this arrangement, equalizer tapsmay be selectively configured, according to system requirements, toprovide either temporal equalization (i.e., pre-tap and/or post-tapequalization) or cross-talk cancellation.

FIG. 6 illustrates a transmit device 151 having circuitry for selectingbetween temporal equalization and cross-talk cancellation data sources.The transmit device 151 includes transmitters 152 ₁ and 152 ₂, each fortransmitting data signals on a respective signal path 122 ₁ and 122 ₂.Respective sources of transmit data values (TX DATA1 and TX DATA2) areprovided from other logic (not shown) within transmit device 154.Although only two transmitters 152 are shown, additional transmittersmay be provided in accordance with the number of signal paths 122 and/orthe number of sources of transmit data values.

Each of the transmitters 152 includes a transmit shift register (124 ₁,124 ₂), output driver (121 ₁, 121 ₂), post-tap data shift register (120₁, 120 ₂) and output driver bank (131 ₁, 131 ₂) that operate generallyas described in reference to FIG. 3. Each transmitter 152 additionallyincludes a tap data source selector (153 ₁, 153 ₂) having one or moremultiplexers for selectively coupling either a local data value (e.g., apre-tap or post-tap data value from corresponding transmit shiftregister 124 or post-tap data shift register 120) or a remote data value(e.g., a primary value supplied from the head of a transmit shiftregister 124 of another transmitter, or a post-tap data value suppliedfrom the post-tap data shift register of another transmitter) to be theequalization tap data source. For example, multiplexer A within tap datasource selector 153 ₁ has a first input coupled to a storage elementwithin post-tap data shift register 120 ₁ and a second input coupled tothe output of transmit shift register 124 ₂, and selects, according to aselect signal SEL_(1A), either a post-tap data value within shiftregister 120 ₁, or the remote primary value output by shift register 124₂ to be the tap data source for an output driver within output driverbank 131 ₁. Multiplexer J within tap data source selector 153 ₁ has afirst input coupled to a storage element within the transmit shiftregister 124 ₁ and a second input coupled to the output of the transmitshift register 124 ₂, and selects, according to a select signalSEL_(1J), either a pre-tap data value within the transmit shift register124 ₁, or the remote primary value to be the data tap source for anoutput driver within output driver bank 131 ₁.

To enable cancellation of crosstalk interference that lasts for morethan a single symbol time, additional multiplexers may be providedwithin the tap data source selectors 153 to select between local datavalues (pre- or post-tap) and remote post-tap data values. For example,multiplexer K within tap data source selector 153 ₁ has a first inputcoupled to receive a pre-tap data value from transmit shift register 124₁ and a second input coupled to receive a remote post-tap data valuefrom post-tap register 120 ₂, and selects between the two inputsaccording to select signal SEL_(1K). Tap data source selector 153 ₂similarly includes one or more multiplexers to select between pre-tap,post-tap and/or cross-talk cancellation data sources for output driverbank 131 ₂. By this arrangement, output drivers within banks 131 mayalternatively be used to generate temporal equalization signals orcross-talk cancellation signals according to system needs.

Although the multiplexers within tap data source selectors 153 ₁ and 153₂ are depicted as two-input multiplexers, multiplexers having more thantwo inputs may alternatively be used. For example, multiplexer A of datasource selector 153 ₁ may include one or more inputs to receive pre-tapdata values from register 124 ₁, one or more inputs to receive post-tapdata values from post-tap register 120 ₁, and/or one or more inputs toreceive cross-talk cancellation data values (i.e., remote primary,pre-tap and/or post-tap values from any number of other transmitters152). In general, each output driver within an output driver bank 131may be sourced by a multiplexer that selects between any number ofpre-tap, post-tap and/or cross-talk cancellation data sources. Also, notall output drivers within output driver banks 131 need be fed bymultiplexers, but rather may be coupled to dedicated tap data sources.

In one embodiment, the select signals, SEL₁ (including signals SEL_(1A),SEL_(1J), SEL_(1K), etc.) and SEL₂, are generated by a configurationcircuit (not shown) within transmit device 151 or elsewhere in asignaling system that includes transmit device 151. The configurationcircuit may be pre-programmed or may be programmed at system start-up,for example, by a controller similar to controller 141 of FIG. 3.

As described in reference FIG. 6, selective-tap transmit-sidepreemphasis may be used to cancel or reduce interference between signalstransmitted in the same direction on neighboring or otherwise proximalsignal lines (i.e., far-end cross-talk). Selective-tap receive-sideequalization may similarly be used to reduce interference betweenoutgoing and incoming transmissions on proximal signal lines;interference referred to herein as near-end cross-talk.

FIG. 7 illustrates transmit and receive devices (118 and 119,respectively) configured to perform near-end cross-talk cancellation.The transmit device 118 includes an output driver 121, transmit shiftregister 124, post-tap data shift register 120, and output driver bank131, all of which operate generally as described above in reference totransmit device 115 of FIG. 3 to enable generation of an equalizedtransmit signal (TX OUT) on signal path 122 ₁. Though not shown in FIG.7, the transmit device 118 may additionally include select circuitry asdescribed in reference to FIG. 6 to enable selection of variousequalization data sources.

The receive device 119 includes a sampling circuit 123, buffer 132, tapselect circuit 137, tap select logic 139 and equalization circuit (e.g.,included within the sampling circuit 123 or implemented as an outputdriver 140) to receive an incoming signal (RX IN) on signal path 122 ₂.As shown in FIG. 7, the incoming signal, RX IN, has a smaller amplitudethan the transmit signal, TX OUT, (e.g., due to transmission losses) andtherefore is particularly susceptible to near-end cross-talkinterference. To counteract cross-talk interference from the TX OUTtransmission, pre-tap, primary and post-tap data values used to generatethe TX OUT signal (i.e., from transmit shift register 124 and post-tapdata shift register 120) are supplied to the tap select circuit 137within the receiver 119. By this arrangement, the tap select logic 139may select, as tap values for the receiver equalization circuit, anycombination of the received data values stored within buffer 132, andthe pre-tap, post-tap and primary data values used to generate the TXOUT signal. As discussed above in reference to FIG. 3, tap select logic139 outputs a control signal to the tap selector 137 to control tap datasource selection according system configuration information. Thus, thepre-tap, post-tap and/or primary data values may be selected with thepolarity necessary to achieve a subtractive effect on the correspondingcross-talk interference (the appropriate polarity being established orcontrolled within the buffer 132 or receiver equalizing circuit),thereby enabling reduction of near-end cross-talk interference. Althoughonly a single tap select circuit 137 is shown in FIG. 7, any number oftap select circuits may be used.

Bi-Directional Signaling

Although a unidirectional signaling system is depicted in FIG. 3,embodiments of the invention are equally applicable in a bidirectionalsignaling system. FIG. 8, for example, illustrates a transceiver device151 that may be coupled to either or both sides of signal path 122, andthat includes both an equalizing transmitter 115 and an equalizingreceiver 116 according to embodiments described herein (transmitters andreceivers according to the cross-talk canceling embodiments described inreference to FIGS. 6 and 7 may also be used). The transceiver device 151additionally includes an application logic circuit 154 to providetransmit data to the equalizing transmitter 115 and to receive sampleddata from the equalizing receiver 116. The application logic circuit 154also outputs an enable signal (ENABLE) to alternately enable thetransmitter 115 to transmit data on the signal path 122 or the receiver116 receive data from the signal path 122.

FIG. 9 illustrates an equalizing transceiver 155 according to anembodiment in which both transmitted and received data values are storedand selectively used to source equalizer taps. The transceiver 155includes a transmit shift register 124, output driver 121, post-tap datashift register 120 and output driver bank 131 (which may include outputdrivers sourced by pre-tap data values, cross-talk cancellation values,or by tap data source selectors as described in reference to FIG. 6),all of which operate generally as described in reference to FIG. 3 tooutput, during a given transmit interval, a primary signal andcorresponding equalization signal onto signal path 122. The transceiveralso includes a sampling circuit 123, buffer circuit 132, tap selector156 and tap select logic 157. The sampling circuit 123 samples datasignals transmitted on signal path 122 (i.e., by a remote transmitter ortransceiver) and stores the corresponding data values in buffer circuit132. The tap selector 156 is coupled to the buffer circuit 132 as wellas the transmit shift register (including the head of the transmit shiftregister which contains the primary data value) and the post-tap datashift register 120, and therefore enables any combination of receiveddata values (i.e., from buffer 132) and pre-tap, primary and/or post-taptransmit data values to be selected as source data taps within anequalizing circuit (i.e., output driver 140 or an equalizing circuitwithin the sampling circuit 123). The tap select logic 157 outputs acontrol signal to the tap selector according system configurationinformation (i.e., information indicative of desired symbol latencies)and the historical state of the enable signal (ENABLE). Thus, dependingon the desired symbol latencies of data taps, and the times at which thetransceiver 155 is transitioned between sending and receiving data(i.e., turnaround times), the tap select logic 157 and tap selector 156operate to select tap values from the transmit shift register 124, datatap shift register 120, and/or buffer circuit 132 in any combination.The selected tap values are then used to source equalizer taps within anequalizing output driver 140 or an equalizing circuit within samplingcircuit 123.

Although the transceiver embodiments described in reference to FIGS. 8and 9 include an enable line to alternately enable transmission orreception of signals, in alternative embodiments, the enable line may beomitted and transmission and reception of signals may occursimultaneously (i.e., simultaneous bi-directional signaling). In such asystem, multilevel signaling may be used to enable an outgoing signal tobe transmitted simultaneously (in effect, superimposed on) an incomingsignal. Because the receive circuit has access to the transmitted datavalues, the receive circuit may subtract the locally transmitted signalfrom an incoming signal to recover only the desired portion (i.e.,remotely transmitted portion) of the incoming signal. In such a system,the locally transmitted signal may produce dispersion- andreflection-type ISI that may be compensated by an equalizing receiverhaving, as an example, the configuration of FIG. 9, but omitting theenable line. In such an embodiment, the transmit shift register 124 andpost-tap data register 120 may be selected by tap select circuit 156 tosource tap data values for equalization of low- and/or high-latencydistortions resulting from the local signal transmission (i.e., byoutput driver 121). Note that the post-tap data register may need to beextended (i.e., have an increased number of entries) to enable reductionof high-latency distortions resulting from the local signaltransmission. The receive circuit tap selections, controlled by tapselect logic 157, may be determined empirically or during run-time, forexample, by using the methods and circuits described below fordetermining equalization tap latencies, weights and polarities.

Data Tap Selection

FIG. 10 illustrates an exemplary buffer 159 that may be used within thereceiver 116 of FIG. 3 and that includes both a serial shift register161 as well as a number (K) of parallel-load registers 165 ₁-165 _(K).At each edge of a receive clock signal, RCLK, a newly sampled data value160 is loaded from sampling circuit 123 into the shift register 161. Theshift register is formed by N storage elements (depicted as flip-flops163 ₁-163 _(N), though latches or other types of storage elements may beused) coupled in daisy chain fashion such that, as the newly sampledvalue 160 is loaded into the first storage element 163 ₁ in the shiftregister 161, the contents of each storage element 163 except the last(163 _(N)) is shifted to the next storage element in the chain inresponse to a receive clock signal (RCLK). Thus, designating the outputof storage element 163 ₁ to have symbol latency i, the symbol latency ofthe input value 160 is i−1, and the symbol latencies of the outputs ofthe remaining storage elements 163 in the shift register 161 are, fromleft to right, i+1, i+2, . . . , and i+(N−1), respectively.

A shift counter 169 (which may be included within or separate frombuffer circuit 159) maintains a count of the number of data valuesshifted into the shift register 161, incrementing the shift count inresponse to each transition of RCLK. In one embodiment, the shiftcounter 169 asserts a load signal 164 (LD) upon reaching a count thatcorresponds to a full shift register, then rolls the shift count back toa starting value. The load signal 164 is routed to strobe inputs ofstorage elements within the parallel-load registers 165, enablingparallel load register 165, to be loaded with the contents of the shiftregister, and enabling each of the parallel-load registers 165 ₂-165_(K) to be loaded with the content of a preceding one of the parallelload registers (i.e., 165 ₂ is loaded with the content of 165 ₁, 165 ₃is loaded with 165 ₂, and so forth). By this arrangement, the symbollatency of a data value stored within any of the parallel-load registers165 is dependent on how many data values have been shifted into theshift register since the last assertion of the load signal 164; ameasure indicated by the shift count. For example, if the shift count is1, indicating that the load signal 164 was asserted at the immediatelypreceding edge of RCLK, then the content of storage element 167 ₁ ofparallel-load register 165 ₁ has a symbol latency of i+1 (i.e., onesymbol time older than the content of storage element 163 ₁ of the shiftregister). When the next value is shifted into the serial shift register161, the contents of the parallel registers 165 remain unchanged,meaning that the latency of each data value stored in the parallelregisters 165 is increased by a symbol time. Thus, the content latency(i.e., latency of a stored value) of a given storage element within oneof parallel registers 165 is dependent upon the value of the shiftcount. Referring to parallel load register 165 ₁, for example, thecontent latency of storage element 167 ₁ is i+SC (SC being the shiftcount), the content latency of storage element 167 ₂ is i+SC+1, and soforth to storage element 167 _(N), which has a content latency ofi+(N−1)+SC. The content latencies of storage elements within theparallel-load registers 165 ₂-165 _(K) are similarly dependent upon theshift clock value, SC, but are increased by N for each parallel loadaway from register 165 ₁. That is, the content latency of the leftmoststorage element within register 165 ₂ is i+N+SC, and the content latencyof the leftmost storage element within register 165 _(K) is i+(K-1)N+SC.The content latencies of the storage elements within registers 165 ₂-165_(K) are incrementally related to the content latency of thecorresponding leftmost storage element in the same manner that thecontent latencies of storage elements 167 ₂-167 _(N) relate to thecontent latency of storage element 167 ₁.

FIG. 11 illustrates, in flow diagram form, a method of selecting a datavalue having symbol latency i+X from the buffer 159 of FIG. 10, i beingthe content latency of storage element 163 ₁. At 175, X is compared withN, the number of storage elements within the shift register 161 andwithin each of the parallel-load registers 165. If X is less than N,then the desired data value is located within the shift register which,after being initially loaded, always contains data values having symbollatencies ranging from i to i+N−1. Thus, if X is less than N, then asshown at 177, the desired value is in the shift register (REG=SR) at bitposition X (BIT=X), where the bit position corresponds to left-to-rightnumbered storage elements.

If X is not less than N, then the desired data value is located at ashift-count-dependent bit position within one of the parallel-loadregisters 165. Thus, if X is less than N+SC (179), the desired datavalue is located within register 165 ₁ at bit position X−SC, asindicated at 181. To understand this result, consider what happens if adata value having a desired symbol latency is initially within therightmost storage element, 167 _(N), within parallel-load register 165₁. As a new value is shifted into the serial shift register 161 and theshift count is incremented, the symbol latency of storage element 167_(N) is increased, and the storage element one position to the left ofstorage element 167 _(N) (i.e., 167 _(N−1)) now contains the data valuehaving the desired symbol latency and is therefore selected to supplythe data value to an equalizer tap.

Returning to FIG. 11, if X is greater than or equal to N+SC, then X iscompared with 2N+SC at 183. If X is less than 2N+SC, then parallel-loadregister 165 ₂ contains the desired tap value at bit position X−N−SC asindicated at 185. The decision flow continues in this manner to 187 atwhich point X is compared with (K−1)N+SC. If X is less than (K−1)N+SC,then parallel-load register 165 _(K−1) contains the desired tap value atposition X−(K−1)N−SC as indicated at 189. Otherwise, X is located withinthe final parallel-load register, 165 _(K) at position X−KN−SC asindicated at 191.

FIG. 12 illustrates an exemplary embodiment of a tap select circuit forselecting a tap value (DATA_(i+X)) from a buffer circuit 210 thatincludes an eight-bit serial shift register 161 and two eight bitparallel-load registers 165 ₁, 165 ₂. For purposes of illustration, itis assumed that the data value in the first (leftmost) storage elementwithin the shift register 161 has a symbol latency of one and that thedead range is four symbol times (i.e., the leftmost four storageelements within the shift register 161 are not used to source tap valuesto the equalizer). By this arrangement, immediately after a parallelload operation, the data values stored in parallel-load register 165 ₁will have symbol latencies ranging from 2-9 symbol times, and the datavalues stored in parallel-load register 165 ₂ will have symbol latenciesranging from 10-17 symbol times. Accordingly, reflections (or otherdistortions) appearing at the receiver input between 5 and 17 symboltimes after the corresponding primary signal may be reduced by selectingdata values having corresponding symbol latencies from the buffercircuit 210 to drive the receive-side equalizer taps (i.e., to be tapdata values). Multiplexers 205, 207 ₁ and 207 ₂ are responsive to loworder bits of a latency value 200 (X[4:0]) to select tap positionswithin the shift register 161, parallel-load register 165 ₁, andparallel-load register 165 ₂. The latency value 200 is additionallysupplied to a select logic circuit 201 which generates a register selectsignal, SEL[1:0], to select one of the three registers 161, 165 ₁ and165 ₂ within the buffer circuit 210 to source the tap data value,DATA_(i+X). It should be noted that, because the range of tap valuesextends over 13 possible symbol times (symbol latencies from 5-17), asmaller, four-bit latency value may alternatively be used to select atap value. As described below, however, using a latency value largeenough to select any of the bit positions within the buffer circuit 210enables the size of the dead range to be adjusted (i.e., programmed)according to application needs.

The least significant two bits of the latency value 200 (i.e., X[1:0])are input to multiplexer 205 to select one of the four selectable datavalues within the serial shift register 161. The least three significantbits of the latency value 200 (i.e., X[2:0]) are input to a subtractcircuit 203 which subtracts the shift count 202 from the three-bitlatency value to produce a tap select value for the parallel-loadregisters 165 ₁, 165 ₂. In one embodiment, for example, the select value200 corresponds to a desired symbol latency as shown in Table 1 below,and the shift count 202 is encoded in a three-bit value, SC[2:0], asshown in Table 2 below. Thus, when the shift count 202 is eight(SC[2:0]=000), and the select value is nine (X[4:0]=01000), the outputof the subtract circuit 203 will be: X[2:0]−SC[2:0]=0; a value thatcorresponds to the leftmost bit position within each of theparallel-load registers 165. This is a desired result as the leftmostbit positions within registers 165 have symbol latencies 9 and 17 whenthe shift count is eight. The multiplexer 209 will generate a selectsignal 204 (SEL[1:0]) to select the data value from register 165 ₁(symbol latency=9) to source the tap data value (this operation isdiscussed below). As a further example, when the shift count 202 is oneand the select value 200 is nine, the output of the subtract circuit 203will be: 000−001=111 (decimal 7); the rightmost bit position within eachof the parallel-load registers 207. Again, this is a desired result asthe rightmost bit positions with registers 165 have symbol latencies 9and 17 when the shift count is one.

TABLE 1 X[4] X[3] X[2] X[1] X[0] Tap Latency 0 0 1 0 0 5 0 0 1 0 1 6 0 01 1 0 7 0 0 1 1 1 8 0 1 0 0 0 9 0 1 1 0 0 10 0 1 1 0 0 11 0 1 1 0 0 12 01 1 0 0 13 0 1 1 0 0 14 0 1 1 0 0 15 0 1 1 0 0 16 1 0 0 0 0 17

TABLE 2 SC[2] SC[1] SC[0] Shift Count 0 0 1 1 0 1 0 2 0 1 1 3 1 0 0 4 11 0 5 1 1 1 6 1 1 1 7 0 0 0 8

FIG. 13 illustrates an exemplary embodiment of the select logic 201 ofFIG. 12. The select logic 201 includes a comparator circuit 215 tocompare the latency select value 200 with N (the size, in bits, of eachof registers within buffer circuit 210), a summing circuit 217 to sumthe shift count 202 with N (thereby generating SC+N), and a comparatorcircuit 219 to compare the latency select value 202 with the output ofthe summing circuit 217. If the latency select value 200 is less than N,the output of comparator 215 will go high, causing inverter 223 to drivethe high order bit of select signal 204 low and OR gate 221 to drive thehigh order bit of the select signal 204 high. That is, SEL[1:0]=01 sothat multiplexer 209 will select the shift register 161 to source thetap data value; the desired result when the select value 200 is lessthan N. If the select value 200 is greater than or equal to N, theoutput of comparator 215 will go low, thereby driving the high order bitof select signal 204 high, and enabling the output of comparator 219 tocontrol, via OR gate 221, the state of the low order bit of selectsignal 204. If the latency select value 200 is less than the output ofsumming circuit 217 (SC+N), the output of comparator 219 will be lowcausing select signal SEL[0] to be low, thereby producing SEL[1:0]=10and selecting parallel-load register 165 ₁ to source the tap data value.If the latency select value 200 is greater than or equal to the outputof the summing circuit 217, the output of comparator 219 will be high,resulting in a select value of SEL[1:0]=11, thereby selectingparallel-load register 165 ₂ to source the tap data value.

Reflecting on the structure of FIG. 13, it should be noted that thesumming circuit and comparators may have numerous implementationsdepending on the size of N and the number of bits used to form thelatency select value 200 and shift count 202. For example, where thelatency select value 200 and shift count 202 have the five- andthree-bit configurations shown, and N is eight, the sum of the shiftcount and N (performed by circuit 217 in FIG. 13) may be formed simplyby including an additional bit in parallel with the three shift countbits, the additional bit forming the most significant bit of theresulting sum (i.e., sum[3]) while SC[2:0] form the less significantthree bits of the sum (i.e., sum[2:0]). As another example, thecomparator 215 may be implemented by a NOR gate having inputs coupled toX[4] and X[3]. By this arrangement, the X<N output will be high only ifboth X[4] and X[3] are low. Numerous other logic circuits may be used toimplement the select logic circuit 201 of FIG. 12 in alternativeembodiments. More generally, specific numbers of bits and registers havebeen described for purpose of example only. Alternative embodiments mayinclude different numbers of registers having various sizes, and latencyselect values and shift count values having different sizes. Also, anycircuit for selecting a data value based on a latency select value mayalternatively be used without departing from the spirit and scope of thepresent invention.

FIG. 14 illustrates a generalized select circuit 230 that may be used toselect Q tap values from the buffer circuit 210 of FIG. 12. The selectcircuit 230 includes a switch matrix 231 and tap select logic 235. Inthe embodiment of FIG. 14, each of the possible tap data sources withinthe buffer circuit 210 (i.e., the rightmost four bits within shiftregister 161 and all the bits within the parallel-load registers 165)are coupled to respective column lines 234 of the switch matrix 231, andeach of the Q tap outputs are coupled to respective row lines 236 of theswitch matrix 231. A switch element 233 is provided at each row-columnintersection to enable the tap data source for the column to beselectively coupled to the tap output for the row. The tap select logic235 outputs a respective one of enable signals E₁-E₂₀ to each column ofswitch elements based on latency selection values X₁-X_(Q) and shiftcount, SC. In the embodiment of FIG. 14, each enable signal includes Qcomponent signals coupled respectively to the Q switch elements within acorresponding column. Thus, if the column 1 data value (i.e., the datavalue stored in shift register position 4), is selected to be the datasource for tap Q, then select signal E₁[Q:1]=100.00. More generally,E_(j)[i]=1 for each column data value, j, to be coupled to a tap output,i. By this arrangement, the Q tap outputs may be selected from among thecomplete range of data values stored within buffer circuit 210. In oneembodiment, the select logic includes combinatorial logic that operatesas described in reference to FIG. 10 to generate each enable signal.Alternatively, a state machine or other processing logic may be used togenerate the enable signals in accordance with the latency selectionvalues and shift count.

FIG. 15 illustrates an embodiment of a switch element 233 that may beused within the switch matrix 231 of FIG. 14. The switch elementincludes a transistor 235 having source and drain terminals coupledbetween the i^(th) row line 236 _(i) (TAP_(i)) and the j^(th) columnline 235 _(j) (DATA_(j)) of the switch matrix, and a gate terminalcoupled to receive the i^(th) component signal of enable signal j (i.e.,E_(j)[i]). Thus, when the E_(j)[i] is high, indicating that i^(th) tapoutput is to be sourced by the data value at the j^(th) position withinthe range of selectable data values, transistor 235 is switched on tocouple the selected data source to the tap output. Other types ofswitching elements may be used in alternative embodiments.

Equalizing Circuits

As discussed above in reference to FIG. 3, the tap values selected bythe tap select logic 139 and select circuit 128 may be used in a numberof different equalizing circuits to counteract distortion events. In oneequalizing receiver embodiment, illustrated in FIG. 16, an equalizingoutput driver 140 is coupled in parallel with the sampling circuit 123to drive an equalizing signal back onto the signal path 122 during eachsymbol reception interval (i.e., symbol time during which a valid symbolis present at the input of the receiver). By this arrangement, latentdistortions arriving at the receiver during a symbol reception intervalmay be canceled (or at least reduced) by operation of the equalizingoutput driver 140.

FIG. 17 illustrates the receive circuit of FIG. 16 in greater detail. Asshown, the sampling circuit 123 may include any number of preamplifiers240 ₁-240 _(N) coupled in series with a sampler 241. The sampler 241 maybe any type of circuit for detecting the level of an input signal,including but not limited to a latching circuit that latches the signallevel in response to a rising or falling clock edge, or an integratingcircuit that integrates the input signal over a finite period of time(e.g., a symbol time or portion of a symbol time). The equalizing outputdriver 140 may be coupled to the signal path 122 (i.e., the input of thefirst preamplifier 240 ₁) or, alternatively, to the output of any of thepreamplifiers 240. Also, as discussed below, the output driver 140 maybe coupled to the sampler 241 to affect the sampling operation.

In one embodiment, the equalizing output driver 140 of FIGS. 15 and 16is clocked by an equalizer clock signal, EQCLK, that is offset from theclock signal used to time the sampling instant and therefore to definethe symbol reception interval (i.e., receive clock signal, RCLK), asnecessary to align edges of the equalizing signal (i.e., statetransitions) with edges of the incoming data signal. This timingrelationship is illustrated by FIG. 18. As shown, the equalizer clocksignal is aligned with edges of the incoming symbol stream so thatequalization values are transmitted onto the signal path concurrentlywith corresponding symbol reception intervals. As discussed below, theequalizer clock signal may be further offset from the receive clocksignal as shown by arrow 245 to account for the time required for theequalization data (i.e., selected tap values) to propagate through theequalizing output driver 140 or other equalizing circuit.

FIG. 19 illustrates a current-sinking output driver 250 that may be usedto implement the equalizing output driver 140 of FIG. 16. The outputdriver includes a plurality of sub-driver circuits 251 ₁-251 _(N), eachsub-driver circuit 251 including a current source 257, clockingtransistor 122 and data tap transistor 253 coupled in series between anoutput node 254 and a reference voltage (ground in this example).Control terminals (e.g., gate terminals) of the data tap transistors 253of the sub-driver circuits 251 are coupled to receive respective datatap values (designated EQD₁-EQD_(N) in FIG. 19) from a select circuit,control terminals of the current sources 257 are coupled to respectivetap weight values, EQW₁-EQW_(N), and control terminals of the clockingtransistors are coupled in common to receive the equalizer clock signal,EQCLK. By this arrangement, when the equalizer clock signal goes high,each of the sub-driver circuits will source a current according to itsrespective tap weight and data tap inputs. For example, referring tosub-driver circuit 251 ₁, if tap data value EQD₁ is low, no current (ornegligible) current will be drawn via output node 254. By contrast, iftap data value EQD₁ is high, then the sub-driver circuit 251 ₁ will drawa current from the output node 254 (and therefore from the signal path122) according to the tap weight, EQW₁. As discussed below, the tapweights provided to the output driver 250 or other equalizing circuitsdescribed herein may be predetermined values, or may be determineddynamically according to the level of the distortions to be reduced.Because the sub-driver circuits 251 are coupled in parallel to theoutput node, the overall equalization signal generated by output driver250 during a given symbol time is the sum of contributions from theindividual sub-driver circuits 251. Note that the output driver 250outputs an equalization signal only when the equalizer clock signal ishigh (i.e., even phases of EQCLK). An additional instance of outputdriver 250 may be provided to output an equalization signal when acomplement equalizer clock signal (i.e., /EQLCK) is high.

FIG. 20 illustrates an embodiment of a push-pull type of sub-drivercircuit 260 that may be used within an equalizing output driver insteadof the pull-down sub-driver circuits 251 described in reference to FIG.19. In the push-pull type of sub-driver circuit 260, current is eithersourced or sunk via the driver output according to the state of the tapdata value, EQD_(i). The sub-driver circuit 260 includes switchingtransistors 263 and 265, and AND gate 261. A first input of the AND gate261 is coupled to receive the tap data value, EQD_(i), and a secondinput of the AND gate 261 is coupled to a clock line to receive theequalizer clock signal, EQCLK. The output of the AND gate 261 is coupledto the gate terminals of transistors 263 and 265 such that, during eachhigh phase of the equalizer clock signal, the tap data value is passedto the gate terminals of transistors 263 and 265 to establish the outputstate of the sub-driver circuit 260. That is, every other half cycle ofthe equalizer clock signal constitutes an output enable interval for thesub-driver circuit 260. If the tap data value, EQD_(i), is high during agiven output enable interval, transistor 265 is switched on, causing thesub-driver circuit 260 to sink current via the output node (OUT_(i)).Conversely, if the tap data value is low during the output enableinterval, transistor 263 is switched on to source current via the outputnode. Also, though not shown in FIG. 20, a pull-down biasing circuit(e.g., current source) may be coupled between the pull-down data taptransistor 265 and ground, and a pull-up biasing circuit may be coupledbetween the pull-up data tap transistor 263 and the supply referencevoltage (e.g., V_(DD)) to enable weighted control of the currentsourcing and sinking strength of the push-pull sub-driver circuit 260.Further, an additional instance of the sub-driver circuit 260 may beprovided with a complement equalizer clock signal (/EQCLK) andcomplement tap data value (/EQD_(i)) being input to AND gate 261 toenable the sub-driver circuit 260 to output an equalizing signal duringthe alternate half cycle of the equalizer clock signal.

FIG. 21 illustrates another embodiment of a sub-driver circuit 275 thatmay be used within an equalizing output driver. The sub-driver circuit275 includes a differential transistor pair 277 having control terminalscoupled to outputs of AND gates 261 ₁, and 261 ₂, respectively. A tapdata value (EQD_(i)) and an equalizer clock signal (EQCLK) are input toAND gate 261 ₁, and a complement of the tap data value (/EQD_(i)) andthe equalizer clock signal are input to AND gate 261 ₂. By thisarrangement, the tap data value and complement tap data value areapplied to respective inputs of the differential pair 277 during everyother half cycle of the equalizer clock signal. Output nodes of thedifferential pair 277 are pulled up through respective resistive loads283 (R), and source terminals of the differential pair are coupled toground via a current source 281. The resistive loads 283 may be, forexample, termination elements coupled to the signal path (not shown)rather than resistive elements included within the sub-driver circuit275. Accordingly, the sub-driver circuit 275 is enabled, during everyother half cycle of the equalizer clock signal, to output a differentialequalizing signal on output nodes OUT_(i) and /OUT_(i) in accordancewith the complementary tap data values, EQD_(i) and /EQD_(i). Acounterpart instance of sub-driver circuit 275 may be provided togenerate a differential equalizing signal during the alternate halfclock cycle of the equalizer clock signal. The current source 281 iscontrolled by the tap weight value, EQW_(i), in the manner described inreference to FIG. 19, though different weighting schemes may be used inalternative embodiments (e.g., using weight-biased pull-up elements inplace of resistive elements 283).

FIG. 22 illustrates an alternative type of equalizing circuit 290 thatmay be used in embodiments of the invention. Instead of driving anequalization signal onto the signal path to affect the signal level ofan incoming signal, equalization is performed in conjunction withpreamplification of the incoming signal, and therefore affects the levelof preamplification applied to the incoming signal. That is, theequalizing circuit 290 affects the preamplified signal level instead ofthe signal level present on the signaling path.

Equalizing circuit 290 includes a differential amplifier 294 formed bydifferential transistor pair 291, biasing current source 292 andresistive loads 293. Differential input signals are supplied to gateterminals of transistor pair 291 such that differentially amplifiedoutput signals are generated on output lines P_(OUT) and /P_(OUT). Inone embodiment, output lines P_(OUT) and /P_(OUT) are coupled to inputterminals of a differential amplifier within a sampling circuit so thatamplifier 294 effectively forms a first stage in a two-stage amplifier(i.e., amplifier 294 is a preamplifier).

Equalizing circuit 290 additionally includes a level shifting circuit296 coupled to the differential amplifier 294 to provide preamplifierequalization. The level shifting circuit 296 includes a pair ofsub-circuits 298 ₁ and 298 ₂ each coupled between a respective one ofthe differential amplifier outputs (P_(OUT) and /P_(OUT)) and a clockingtransistor 299. Each of the signal subcircuits 298 includes a respectiveplurality of data tap transistors (295 ₁-295 _(N) and 297 ₁-297 _(N))coupled in parallel between the differential amplifier output and theclocking transistor 299. The control terminals of the data taptransistors 295 ₁-295 _(N) are coupled to receive the selected data tapvalues, EQD₁-EQD_(N), respectively, and the control terminals of thedata tap transistors 297 ₁-297 _(N) are similarly coupled to receivecomplement versions of the selected data tap values, /EQD1-/EQDN. In oneembodiment, each of the data tap transistors 295 is sized (e.g., bywidth-length ratio) to achieve a respective tap weight EQW_(N)-EQW₁. Bythis arrangement, each data tap value may be coupled to the controlterminal of a selected one of the data tap transistors 295 according tothe desired tap weight. The transistors 297 are similarly weighted andtherefore allow coupling of the complement data tap values according todesired tap weights. The weights of the individual data tap transistors295 (and 297) may be incrementally related (i.e., EQW₁=EQW₂+K=EQW₃+2K .. . , where K is a constant), exponentially related (i.e.,EQW₁=EQW₂*K=EQW₃*K² . . . ) or may have any other desired relationship(including having the same weight values or including subsets of weightvalues that are the same).

The clocking transistor 299 is switched on during every other half cycleof the equalizer clock signal to enable the operation of the subcircuits298. The subcircuits 298 operate to increase or decrease the differencebetween the preamplified output signals (or even change the polarity ofthe difference) by drawing more current from one of the preamplifieroutput lines (P_(OUT) or /P_(OUT)) than the other in accordance with theselected data tap values. Thus, the subcircuits 298 act todifferentially shift the level of the preamplified output signalgenerated by differential amplifier 294. An additional instance of theequalizing circuit 290 may be provided to enable preamplifierequalization during the alternate half cycle of the equalizer clocksignal.

FIG. 23 illustrates an alternative level shifting circuit 305 that maybe substituted for circuit 296 of FIG. 22. In circuit 305, differentialpairs of data tap transistors 307 ₁-307 _(N) are coupled to output linesPOUT and /POUT in the same manner as in circuit 296, but instead ofsizing the data tap transistors to achieve tap weighting, tap weightedcurrent sources 311 ₁-311 _(N) are coupled in series with thedifferential pairs of data tap transistors 307 ₁-307 _(N), respectively.For example, current source 311 ₁ is controlled by (i.e., draws a biascurrent according to) weight value EQW₁ and is coupled via clockingtransistors 309 ₁ to data tap transistors 307 ₁. Similarly, currentsource 311 ₂ is controlled by weight value EQW₂ and is coupled viaclocking transistors 309 ₂ to transistors 307 ₂, and so forth. By thisarrangement, the weight values EQW₁-EQW_(N) may be configured (e.g., viarun-time calibration or production time programming) as necessary toestablish a desired equalizing signal contribution from eachdifferential pair of data tap values 307. An additional instance of theequalizing circuit 290 may be provided to enable preamplifierequalization during the alternate half cycle of the equalizer clocksignal (i.e., by driving clocking transistors 309 with complementequalizing clock, /EQCLK).

FIG. 24 illustrates another type of equalizing circuit 320 that may beused in embodiments of the invention. Instead of driving an equalizationsignal onto the signal path to affect the signal level of an incomingsignal, or affecting the preamplified signal level, a level shiftingcircuit 330 is coupled to low impedance inputs of a differentialsampling circuit 328, and is used to affect the level of the inputsignal before the sampled signal is captured. The sampling circuitincludes differential transistor pair 329 to precharge input nodesS_(IN) and /S_(IN) according to the state of a differential input (e.g.,the output of a preamplifier 294 of FIG. 22, or a differential datasignal), during a first half cycle of the receive clock (which enablesclocking transistor 331). During a second half cycle of the receiveclock signal, transistors 321 and 325 are switched on by the low-goingreceive clock signal, thereby enabling a cross-coupled latch formed bytransistors 322, 323, 325 and 326 to latch the state of the prechargedsignal levels on nodes S_(IN) and /S_(IN).

The level shifting circuit 330 is similar to the circuit 296 of FIG. 22except that clocking transistor 341 is enabled by the receive clocksignal (RCLK) instead of the equalizer clock signal, the equalizer clocksignal being used to switch on switching transistors 335 ₁-335 _(N) and339 ₁-339 _(N) during every other half cycle. Data tap transistors 333₁-333 _(N), which are controlled by respective tap data valuesEQD₁-EQD_(N), are coupled in series with the switching transistors 335₁-335 _(N), respectively. Similarly, data tap transistors 337 ₁-337 _(N)are coupled in series with switching transistors 339 ₁-339 _(N) and arecontrolled by respective complement tap data values /EQD₁-/EQD_(N). Inone embodiment, the data tap transistors 333, 337 and switchingtransistors 335, 339 are sized to provide different current drawsaccording to predetermined weights, EQW1-EQWN, thereby permittingdifferent data taps to make different level-shifting contributions. Inone embodiment, for example, the switching transistors 335 and 339 arebinary weighted such that, when switched on, the current draw throughtransistor pair 333 _(N)/335 _(N) is 2^(N−1) times the current throughtransistor pair 333 ₁/335 ₁ (and the current draw through transistorpair 337 _(N)/339 _(N) is 2^(N−1) times the current through transistorpair 337 ₁/339 ₁. Other weighting schemes may also be used including,without limitation, thermometer coding of high-gain transistor pairs,linear weighting schemes, or any combination of exponential (e.g.,binary), linear and thermometer coded weightings.

In one embodiment, the equalizer clock is phase advanced relative to thereceive clock signal such that transistors 337 and 339 are switched onin advance of clocking transistor 341. By this arrangement, transistors333 and 337 are poised to shift the level of the sampling circuit inputnodes, S_(IN) and /S_(IN), when the receive clock signal goes high.Thus, when the receive clock signal goes high, sampling circuit inputnodes S_(IN) and /S_(IN) are differentially discharged according to thetap data values EQD₁-EQD_(N), /EQD₁-/EQD_(N) and the respective weightsof transistors 333 and 337. Consequently, the signal levels at the inputnodes, S_(IN) and /S_(IN), of sampling circuit 328 are differentiallyshifted by the level shifting circuit 330 to reduce static offsets inthe incoming data signal (applied to control terminals of differentialpair 329) caused by reflections or other distortions.

FIG. 25 illustrates an alternative level shifting circuit 342 that maybe substituted for circuit 330 of FIG. 24. The level shifting circuit342 includes data tap transistors 333, 33 and equalizer-clock-enabledswitching transistors 335, 339 coupled as described in reference to FIG.24. However, rather than being coupled to a clocking transistor 335, thesource terminals of transistors 335 ₁-335 _(N) are coupled to ground viacapacitive elements 334 ₁-334 _(N), respectively, and the sourceterminals of transistors 339 ₁-339 _(N) are similarly coupled to groundvia capacitive elements 338 ₁-338 _(N). By this arrangement, respectivevariable capacitances are coupled to the low impedance inputs, S_(IN)and /S_(IN), of the sampling circuit 328 according to the states of thetap data values EQD₁-EQD_(N) and complement data tap values EQD₁-EQD_(N)applied to the inputs of data tap transistors 333 and 337. Thus,different levels of capacitance are added to the sampling circuit inputnodes, S_(IN) and /S_(IN), according to the tap data values, effectivelychanging the discharge rates of the input nodes and therefore affectingthe precharged signal level at the input nodes as desired to reducesignal distortions. In the embodiment of FIG. 25, the data taptransistors 333, 337 and switching transistors 335, 339 have uniformsizes (i.e., uniform weighting), and the capacitive elements 334, 338have weighted capacitive values to permit a broad range of capacitancesto be coupled to the input nodes of sampling circuit 328. For example,in one embodiment, the capacitive elements 334 are implemented bysource-to-drain coupled transistors and are binary weighted (e.g., byadjusting transistor width-length ratios) such that capacitive element3352 has twice the capacitance of capacitive element 335 ₁, andcapacitive element 335 _(N) has 2^(N−1) times the capacitance ofcapacitive element 335 ₁. Other weighting relationships (e.g.,thermometer coding, linear, uniform, etc.) may also be used. Also, thedata tap transistors 333, 337 and/or switching transistors 335, 339 maybe weighted in alternative embodiments instead of (or in addition to)the capacitive elements 334, 338.

High Speed Tap Selector

As discussed above in reference to FIG. 3, an incoming data signal mayinclude two symbols per receive clock cycle (sometimes referred to as a“double data rate” signal), and each symbol may have more than twopossible states (i.e., may have a signal level falling within more thantwo distinct ranges of signals). Also, the receive clock frequency maybe so high that by the time a sampled data value is loaded into thebuffer circuit 132, the data value already has a latency of severalsymbol times. All these factors present challenges to the buffering andselection of tap values described in reference to FIG. 3.

FIG. 26 illustrates an equalizing receiver 350 according to anembodiment of the invention. The receiver 350 includes a double datarate sampling circuit 351, shift register 353, select circuit 355 andequalizing output driver 357. The sampling circuit 351 includes a pairof sub-circuits 3611 and 3612 to sample the incoming data signal inresponse to rising edges in the receive clock (RCLK) and complementreceive clock (/RCLK), respectively. Falling clock edges mayalternatively be used to time the sampling instant. Data samplescaptured in response to edges of the receive clock are referred toherein as even phase data, and data samples captured in response toedges of the complement receive clock are referred to as odd phase data.Thus, sampling circuit 351 outputs even phase data (EVEN IN) and oddphase data (ODD IN) to the shift register 353 via signal lines 362 ₁ and362 ₂, respectively. The even and odd phase data values are storedwithin the shift register to provide a source of selectable tap valuesto the select circuit 355. In the embodiment of FIG. 26, the dead rangeis assumed to be five symbol latencies (other dead ranges may be used)such that data values D_(T+5)-D_(T+X) are provided to the select circuit355, subscript T+5 indicating a latency of five symbol times relative tosampling instant, T. The select circuit 355 includes N tap selectors,365 ₁-365 _(N), that select from among the plurality of data valuesstored within the shift register 353 and output a selected tap datavalue to a respective one of N output sub-drivers 369 ₁-369 _(N) withinthe equalizing output driver 357. Each of the output sub-drivers 369, inturn, drives a component equalizing signal onto the signal path 122. Inalternative embodiments, the equalizing output driver 357 may bereplaced by an equalizing circuit that operates within a preamplifiercircuit (not shown in FIG. 26) or sampling circuit 351 as describedabove in reference to FIGS. 17-20.

FIG. 27 illustrates the shift register 353 and one of the tap selectors365 of FIG. 26 according to more specific embodiments. The shiftregister includes a pair of shift sub-circuits 383 ₁ and 383 ₂ to storeeven phase data and odd phase data, respectively. In one embodiment,each of the shift sub-circuits 383 includes a number of storage elements381 (e.g., latches) coupled in a daisy chain configuration (i.e., outputto input) to enable an input data value to be shifted progressively froma first (i.e., leftmost) storage element 381 in the chain to a last(rightmost) storage element 381 in the chain. Each of the shiftsub-circuits 383 is responsive to the receive clock and complementreceive clock signals such that the contents of each shift sub-circuit383 is shifted during each half clock cycle of the receive clock signal.Thus, assuming that a stream of incoming symbols includes the datasequence A, B, C, D, etc., then even phase data values A, C, E, G, I,etc. will be shifted into shift sub-circuit 383 ₁ and odd phase datavalues B, D, F and H will be shifted in to shift sub-circuit 383 ₂.Because the contents of the shift sub-circuit 383 ₁ are shifted twiceper even phase data reception, two instances of each even phase datavalue will be stored in the shift sub-circuit 383 ₁. The second instanceof each even phase data value stored in shift sub-circuit 383 ₁ isdesignated by a prime (i.e., ′) in FIG. 27 to indicate that the datavalue was loaded synchronously with the loading of a newly received oddphase data value into shift sub-circuit 383 ₂. Similarly, two instancesof each odd phase data value are stored in the shift sub-circuit 383 ₂,with the second instance of the odd phase data value being designated bya prime to indicate that the data value was loaded synchronously withthe loading of a newly received even phase data value into shiftsub-circuit 383 ₁. Thus, from the perspective of the tap selector 365,the shift sub-circuits 383 collectively contain a sequence of datavalues, A′, B, C′, D, E′ F, G′, H, that may be used to generate oddphase equalizing signals (i.e., driving an equalizing signal onto thesignal path or affecting signal levels within a preamplifier or samplingcircuit during odd phase symbol reception), and a sequence of datavalues, B′, C, D′, E, F′, G, H′, I, that may be used to generate evenphase equalizing signals. Accordingly, the outputs of each of thestorage elements 381 within shift sub-circuit 383 ₁ are coupled torespective inputs of an even tap data select circuit 387 ₁ within thetap selector 365, and the outputs of each of the storage elements 381within the shift sub-circuit 383 ₂ are coupled to respective inputs ofan odd tap data select circuit 387 ₂ within the tap selector 365. Theeven and odd tap data select circuits 387 are responsive to a selectsignal, S[2:0], to output selected tap data values from the even and oddphases sequences of data values, respectively. The select signal may begenerated, for example, by tap the select logic 139 described inreference to FIG. 3.

The output of the even tap data select circuit is clocked into aflip-flop 391 ₁ (or other storage element) at the rising edge of thereceive clock signal (RCLK) so that, at any given time, the output offlip-flop 391 ₁ is delayed by two symbol times relative to the mostlatent data value supplied to the even tap data select circuit 387 ₁.Similarly, the output of the odd tap data select circuit 387 ₂ isclocked into a flip-flop 391 ₂ (or other storage element) so that, atany given time, the output of flip-flop 391 ₂ is delayed by two symboltimes relative to the most latent data value supplied to the odd tapdata select circuit 387 ₂. Thus, the flip-flops 391 effectively increasethe latency of selected even and odd data tap values by two symboltimes. Select circuits 393 ₁ and 393 ₂ are provided to extend theoverall latency range of the even and odd data tap selections within tapselector 365 by allowing selection of tap data directly from the evenand odd data inputs to the shift register 353 (i.e., EVEN IN and ODD IN)or from the outputs of flip-flops 391. Select bit S[3] is provided(e.g., by the tap select logic 139 of FIG. 3) to select between the fastpath data (i.e., connections 384 ₁ and 384 ₂ to the inputs of the subshift circuits 383) and the selected data values stored in flip-flops391. Flip-flops 395 ₁ and 395 ₂ (or other storage elements) are providedto synchronize the outputs of multiplexers 393 ₁ and 393 ₂ with thereceive clock and complement receive clock, respectively. Thus, even andodd data tap values, ETD and OTD, each having a range of latenciesaccording to the depth of the shift sub-circuits 383 and the number offast path taps (of which signal lines 384 ₁ and 384 ₂ are examples) areoutput to the equalizing circuit (not shown in FIG. 27) to enable evenand odd phase equalization of an incoming signal.

FIG. 28 illustrates an equalizing receiver 405 for receiving a doubledata rate, multilevel input signal according to an embodiment of theinvention. The receiver 405 includes a sampling circuit 407, shiftregister 411, select circuit 421 and equalizing output driver 427. Thesampling circuit includes even and odd phase sampling sub-circuits 409 ₁and 409 ₂ to capture even and odd phase samples of the incomingmultilevel data signal and to generate a multi-bit output indicative ofthe sampled signal level. For example, in one embodiment, the incomingdata signal has one of four possible signal levels, each level beingdefined by a distinct range of voltages. In the specific embodimentdepicted in FIG. 28, each sample is resolved (i.e., by samplingsub-circuits 409) to a thermometer code in which bits A, B, and C havevalues according to which of four voltage ranges the sampled signallevel falls within. Referring to FIG. 29, for example, bits A, B and Care set according to the following relationships between the sampledsignal, V_(S), and high, middle and low threshold voltage levels:

TABLE 3 Sampled Signal Level, V_(S) C B A V_(S) > T_(H) 0 0 0 T_(H) >V_(S) > T_(M) 0 0 1 T_(M) > V_(S) > T_(L) 0 1 1 T_(L) > V_(S) 1 1 1

Other encoding schemes may be used in alternative embodiments. Also,more or fewer threshold levels (and therefore signal ranges) may beused, and current levels may be used to indicate signal level instead ofvoltage levels.

Once the sampled signal is resolved to a pattern of binary bits, A, Band C (or some other number of bits), each of the bits is input to arespective one of shift registers 413 _(A)-413 _(C) and used to source atap value for selection by a respective set of select circuits 422_(A)-422 _(C) (each select circuit including N tap select selectors 423₁-423 _(N)). Each of the shift registers 413 and select circuits 422operates generally as described in reference to FIGS. 21 and 22 togenerate a set of selected tap values, 424 _(A)-424 _(C). Correspondingtap values from within each set 424 are provided to a respective one ofoutput sub-drivers 429 ₁-429 _(N) within equalizing output driver 427,where they are used to generate a multi-level equalization signal. Forexample, the tap values output by tap selector 423 ₁ within each of theselect circuits 422 are input to output sub-driver 429 ₁ of theequalizing output driver 427.

Equalization Clock Signal Generation

As discussed briefly in reference to FIG. 18, it is desirable for theequalization signal generated by a receive-side equalizing output driverto be driven onto the signal path in phase alignment with data eyes inthe incoming data signal. While the receive clock (or complement receiveclock) may be used to clock the equalizing output driver (or preamp orsampling circuit equalizer), propagation delay through the equalizingdriver tends to become significant in high frequency systems, producingundesired timing offset between the incoming data signal and theequalization signal. In one embodiment, clock data recovery circuitrywithin an equalizing receiver is used to generate an equalization clocksignal (EQCLK) that is phase advanced relative to the receive clocksignal according to the propagation delay (i.e., clock-to-Q) of anequalizing output driver. By this timing arrangement, illustrated inFIG. 30, the equalizing output driver outputs an equalization signalhaving the desired phase relation with the incoming data signal. Asshown, by advancing the equalization clock relative to the receive clockaccording to the clock-to-Q delay of the equalizing output driver, adesired phase relationship between the incoming data signal (RX DATA)and equalization signal (EQ DATA) is achieved. Note that, in theexemplary diagram of FIG. 30, the equalization data tap is assumed tohave a symbol latency of five symbol times, such that an equalizationsignal based on received symbol A is transmitted by the equalizingoutput driver during the reception interval for symbol F.

FIG. 31 illustrates an embodiment of an equalizing receiver 450 thatgenerates receive and equalization clock signals having the phaserelationship shown in FIG. 30. The receiver 450 includes a samplingcircuit 451, shift register 453, clock-data-recovery (CDR) circuit 457,application logic 455, tap data selector 461, signal generator 462,equalizer clock generator 459, and equalization data source selector463. An incoming data signal (DATA) on signal path 122 is sampled by thesampling circuit 451 in response to a receive clock signal (RCLK). Thesamples are output to the shift register 453 where they are stored forparallel output to the application logic 455 and the CDR circuit 457. Inthe embodiment of FIG. 31, the receive clock signal includes multiplecomponent clock signals including a data clock signal and its complementfor capturing even and odd phase data samples, and an edge clock signaland complement edge clock signal for capturing edge samples (i.e.,transitions of the data signal between successive data eyes). The dataand edge samples are shifted into the shift register 453 and thensupplied as parallel words (i.e., a data word and an edge word) to aphase control circuit 467 within the CDR circuit 457. The phase controlcircuit 467 compares adjacent data samples (i.e., successively receiveddata samples) within the data word to determine when data signaltransitions have taken place, then compares an intervening edge samplewith the preceding data sample (or succeeding data sample) to determinewhether the edge sample matches the preceding data sample or succeedingdata sample. If the edge sample matches the data sample that precededthe data signal transition, then the edge clock is deemed to be earlyrelative to the data signal transition. Conversely, if the edge samplematches the data sample that succeeds the data signal transition, thenthe edge clock is deemed to be late relative to the data signaltransition. Depending on whether a majority of such early/latedeterminations indicate an early or late edge clock (i.e., there aremultiple such determinations due to the fact that each edge word/dataword pair includes a sequence of edge and data samples), the phasecontrol circuit 467 asserts an up signal (UP) or down signal (DN). Ifthere is no early/late majority, neither the up signal nor the downsignal is asserted. So long as a calibration signal 474 (CAL) from theapplication logic 455 remains deasserted, the up and down signals, whenasserted, pass through AND gates 468 ₁ and 468 ₂, respectively, toup/down inputs of mix logic 471.

The mix logic circuit 471 receives a set of phase vectors 472 (i.e.,clock signals) from a reference loop circuit 470. The phase vectors haveincrementally offset phase angles within a cycle of a reference clocksignal (REF CLK). For example, in one embodiment, the reference loopoutputs a set of eight phase vectors that are offset from one another by45 degrees (i.e., choosing an arbitrary one of the phase vectors to havea zero degree angle, the remaining seven phase vectors have phase anglesof 45, 90, 135, 180, 225, 270 and 315 degrees). The mix logic 471maintains a phase count value which includes a vector select componentto select a phase-adjacent pair of the phase vectors (i.e., phasevectors that bound a phase angle equal to 360°/N, where N is the totalnumber of phase vectors), and an interpolation component (INT) which isoutput to a mixer circuit 473 along with the selected pair of phasevectors (V1, V2). The mixer circuit mixes the selected pair of phasevectors according to the interpolation component of the phase count togenerate complementary edge clock signals and complementary data clocksignals that collectively constitute the receive clock signal.

The mix logic 471 increments and decrements the phase count value inresponse to assertion of the up and down signals, respectively, therebyshifting the interpolation of the selected pair of phase vectors (or, ifa phase vector boundary is crossed, selecting a new pair of phasevectors) to incrementally retard or advance the phase of the receiveclock signal. For example, when the phase control logic 467 determinesthat the edge clock leads the data transition and asserts the up signal,the mix logic 471 increments the phase count, thereby incrementing theinterpolation component of the count and causing the mixer toincrementally increase the phase offset (retard the phase) of thereceive clock signal. At some point, the phase control signal outputbegins to dither between assertion of the up signal and the down signal,indicating that edge clock components of the receive clock signal havebecome phase aligned with the edges in the incoming data signal.

The equalizer clock generator 459 receives the phase vectors 472 fromthe reference loop 470 and includes mix logic 481 and an equalizer clockmixer 483 that operate in the same manner as the mix logic 471 andreceive clock mixer 473 within the CDR circuit 457. That is, the mixlogic 481 maintains a phase count value that is incrementally adjustedup or down in response to the up and down signals from the phase controlcircuit 467. The mix logic selects a phase-adjacent pair of phasevectors 472 based on a vector select component of the phase count,outputs the selected vectors (V1, V2) and interpolation component of thephase count (INT) to the equalizer clock mixer 483. The equalizer clockmixer 483 mixes the selected vectors in accordance with theinterpolation component of the phase count to generate the equalizerclock signal, EQCLK. The equalizer clock signal, which may includecomplementary component clock signals, is output to the equalizingoutput driver 465 (or other type of equalization circuit as describedabove) to time the output of equalizing signals onto signal path 122.

The equalizer data source selector 463 is responsive to the calibrationsignal 474 to select either the tap selector 461 (which operates asdescribed above to select data tap values from the shift register 453and/or one or more parallel registers) or the signal generator 462 thatoutputs clock pattern 10101010 (e.g., a bi-stable storage element thattoggles between states in response to each EQCLK transition). When thecalibration signal 474 is low, the equalization data source selector 463selects the tap selector 461 to supply selected data values to theequalizing output driver 465. When the calibration signal 474 is high,the receiver 450 enters a calibration mode in which the signal generator462 is selected to supply the clock pattern to the equalizing outputdriver 465. Also, in calibration mode, the high state of the calibrationsignal 474 disables AND gates 468 ₁ and 468 ₂ from passing the up anddown signals to the mix logic 471. Thus, the phase count within the CDRcircuit remains unchanged in calibration mode, while up and down signalsgenerated by the phase control circuit 467 are used to increment anddecrement the phase count value within the mix logic 481. In oneembodiment, no signals are transmitted on the signal path 122 while thereceiver 450 is in calibration mode, so that the only signal present atthe input of the sampling circuit 451 is the clock pattern output by theequalizing output driver 465. By this arrangement, edge and data samplescorresponding to the clock pattern are captured in the shift register453 and supplied to the phase control circuit 467 to determine whetherthe receive clock signal (RCLK) is early or late relative to the clockpattern samples. Accordingly, the phase control circuit 467 will assertan up or down signal (as the case may be) to adjust the phase of thereceive clock signal relative to the incoming data stream. Because thereceive clock phase is effectively locked, however (i.e., by operationof the AND gates 468), only the phase count within the equalizationclock generator will be adjusted. Thus, the normal-mode CDR operation iseffectively carried out in reverse while the receiver 450 is incalibration mode. Instead of shifting the phase of the receive clocksignal to achieve alignment with transitions in the incoming datasignal, the phase of the equalizer clock signal is shifted to aligntransitions in the incoming data signal (i.e., the clock pattern outputby the equalizing output driver) with the receive clock signal. By thisoperation, the equalizer clock signal is advanced relative to an edgeclock component of the receive clock signal by a time substantiallyequal to the clock-to-Q delay of the equalizing output driver 465. Thus,the overall effect of the calibration mode operation is to advance thephase of the equalization clock according to the clock-to-Q time of theequalizing output driver as shown in FIG. 30. In this way, theequalizing output driver 465 drives an equalizing signal onto the signalpath 122 in phase alignment with the incoming data signal.

In one embodiment, the calibration signal 474 is asserted for a timeinterval previously determined to be sufficient to achieve phasealignment between transitions in the transmitted clock pattern and theedge clock component of the receive clock signal. Alternatively, the upand down signals generated by the phase control circuit may be monitoredin the calibration mode to determine when the up and down signals beginto alternate, thereby indicating that the desired phase alignment hasbeen obtained. In either case, after phase alignment has been obtained,the calibration signal is deasserted to enable normal operation of thereceive circuit. At this point, the CDR circuit returns to adjusting thephase count within mix logic 471 in response to the up and down signalsfrom the phase control circuit 467. Because the mix logic 481 within theequalizer clock generator 459 continues to respond to the same up anddown signals, the phase offset between the equalizer clock signal andthe receive clock signal (i.e., the phase offset established in thecalibration mode) is maintained as the phases of the two clocks areadjusted. Thus, in normal-mode operation, the equalizer clock signal andreceive clock signal retain the phase offset established in calibrationmode, but otherwise track one another.

It should be noted that signal patterns other than the clock pattern1010101 may be generated by the signal generator 462 and used to achievethe desired phase relationship between the equalizer clock signal andthe receive clock signal. For example, the signal generator may beimplemented by a pseudo random bit sequence (PRBS) generator thatgenerates a pseudo random bit sequence. More generally, any signalgenerator, random or otherwise, that generates a sequence of valueshaving a sufficient transition density (i.e., transitions per unit time)to enable phase locking in the equalizing receiver 450 (i.e., phaselocking between transitions in the waveform output by output driver 465and the receive clock signal) may be used to implement signal generator462.

Determination of Equalization Tap Latencies, Weights and Polarities

Referring again to FIG. 3, tap selection logic may be implemented in anumber of different ways. In one embodiment, for example, the tap selectlogic 139 includes a configuration circuit that may be programmed withconfiguration information that specifies the tap data sources to beselected by select circuit 128. The configuration circuit may include anonvolatile memory, fusible circuit, etc. that is programmed atproduction time according to the symbol latency, amplitude and polarityof empirically observed (or analytically determined) distortions.Alternatively, the configuration circuit may include memory (volatile ornonvolatile) which is initialized with predetermined configurationinformation during system startup. In yet another embodiment, referredto herein as a self-calibrating embodiment, a signaling system includescircuitry to automatically determine the symbol latency, amplitude andpolarity of distortions on the signaling path between a transmitter andreceiver, and to program a configuration circuit within the tap selectlogic with configuration information that indicates the tap data sourcesto be selected by a select circuit and the tap weights and polarities tobe applied by an equalization circuit.

In one self-calibrating embodiment of the invention, a technique calledembedded scoping is used to determine the symbol latency, amplitude andpolarity of signal path distortions. The symbol latency of a givendistortion, once known, is used to select one or more tap data valueshaving corresponding symbol latencies, and the distortion amplitude andpolarity are used to determine the weight and polarity to be applied tothe selected tap data value in generating an equalization response.Also, the symbol latency of a given distortion may be used to determinewhether to counteract the distortion through transmitter preemphasis orreceiver equalization (or both), and the overall range of symbollatencies for detected distortions may be used to determine anappropriate dead range for the signaling system.

Embedded scoping involves iteratively receiving a sequence of symbols ina receiver and comparing the received symbol sequence with a localgeneration of the sequence to confirm error-free reception. With eachreceive-and-confirm iteration, a threshold voltage used to distinguishbetween symbol values in the incoming signal is offset from a calibratedlevel by a progressively larger amount until a symbol in the sequence nolonger matches the expected value. The threshold voltage offset at whichthe failure occurs is referred to herein as a pass/fail offset andrepresents a measure of the signal level at the sampling instant atwhich the failure occurred. Thus, by sweeping the threshold voltagethrough a range of threshold voltages until the pass/fail offsets foreach symbol in the symbol sequence have been detected, a sample plot forthe incoming signal may be developed. Further, by sweeping the receiveclock signal through an incremental sequence of phase offsets, anddetermining the pass/fail offset at each phase offset, a complete traceof the incoming signal may be generated. Also, the granularity and startstop points of the phase offsets and/or threshold voltage steps may becontrolled (e.g., by configuring a programmable circuit or register) toenable the waveform trace to be constrained to selected points ofinterest in the incoming signal (e.g., ±N° from an intended samplinginstant, N representing a sweep angle).

FIG. 32 illustrates the use of embedded scoping to generate a time-basedtrace 490 of an incoming data signal 486. The range of threshold voltageoffsets over which the incoming signal 486 is sampled is indicated byV_(T), and the range of phase offsets at which the signal is sampled isindicated by φ. Each sample point within the sweep is indicated by arespective dot within a grid of sample points 480. Note that the sweepmay be obtained by stepping the voltage threshold through the range ofV_(T) values for each value of φ, or, alternatively, by stepping theclock phase through the range of φ values for each value of V_(T).

Still referring to FIG. 32, 488 indicates a pair of samples for which apass/fail condition is detected. A corresponding pass/fail offset (PFO)is determined according to the difference between the calibrated V_(T)level (V_(T)(CAL)) and the average of the V_(T) offsets between the passand fail samples, and recorded as a measure of the incoming signal. Thatis, the pass/fail offset may be used to establish a data point withinthe trace 490 as shown. After sweeping through all the sample pointswithin the grid 480 (which sweep may be repeated numerous times toobtain an average and to discard statistical outliers), a measure of theincoming signal is obtained as illustrated graphically by the trace 490.

Embedded scoping has a number of benefits over traditional signalmeasurement techniques. First, because the technique is non-invasive(i.e., no probe contact), the electrical characteristics of the systemunder test are unaltered, thereby yielding potentially more accurateresults. Also, the trace is generated from the perspective of thereceive circuit itself, meaning that any non-ideal characteristics ofthe receive circuit are accounted for in the resulting signal traceinformation. Finally, because all components needed for embedded scopingmay be included within a finished signaling system, embedded scoping maybe used to perform numerous run-time analyses, including determining thelatency and amplitude of reflections and other distortions within thesignaling system.

FIG. 33 illustrates a signaling system 500 according to an embodiment ofthe invention. The signaling system 500 includes a receive device 509and transmit device 501 that employ embedded scoping to determineequalizer tap selections, tap weights and tap polarities. The transmitdevice 501 includes a pattern generator 503, data selector 505,equalizing transmitter 507 and application logic 502. The applicationlogic 502 performs the core function of the transmitting device (e.g.,signal processing, instruction processing, routing control, or any otherfunction) and provides transmit data (TX DATA) to a first input of thedata selector 505. During normal operation, the application logic 502outputs a logic low scope signal 506 (SCOPE) to the data selector 505 toselect the transmit data to be passed to the equalizing transmitter 507for transmission to the receive device 509 via signal path 122 (whichmay include or be connected to numerous sources of discontinuity such asconnectors, vias, stubs, etc.). During a scoping mode of operation, theapplication logic 502 drives the scope signal 506 high to enable ascoping mode of operation within the transmit circuit 501. In thescoping mode, the data selector 505 selects a repeating single-symbolpulse sequence (e.g., a test signal such as: 00100 . . . 00100 . . .00100 . . . ) generated by the pattern generator 503 to be transmittedto the receive device 509. The receive device 509 includes an equalizingreceiver 510 to receive the incoming data signal, a pattern register 511to store a local version of the single-symbol pulse sequence, amultiplexer 512 to enable the pattern register 511 to be switchedbetween load and barrel-shifting modes, a XOR gate 513 to compare thereceived data sequence with the locally generated sequence, andapplication logic 515 (or other logic) to generate a clock adjust signal(CLK ADJ) and threshold voltage adjust signal (THRESH ADJ) to sweep thereceive clock and threshold voltage used within the equalizing receiverthrough their scoping ranges. The application logic 515 additionallybuilds a trace record (i.e., data indicative of the incoming datasequence) based on the output of XOR gate 513.

When the receive device 509 is in a scoping mode of operation, themultiplexer 512 is initially set to load the pattern register 511 withthe output of the equalizing receiver 510. After a desired sequence ofdata (e.g., the single-symbol pulse sequence) is shifted into thepattern register 511, the multiplexer 512 is set to enable thebarrel-shifting mode of the pattern register 513. That is, themultiplexer 512 selects the output of the pattern register 511 to be fedback to the input of the pattern register 511 so that the contents ofthe pattern register 511 are continuously rotated through the patternregister 511 (i.e., a barrel shifting operation). By this arrangement,the data sequence loaded into the pattern register 511 is repeatedlyoutput, bit by bit, to a first input of the XOR gate 513. The datasequence received by the equalizing receiver is input to a second inputof the XOR gate 513 so that the received data sequence is compared, bitby bit, with the data sequence stored within the pattern register 511.By selecting the length of the repeatedly transmitted data sequence tomatch the storage size of the pattern register 511, the pattern registercontents are repeatedly compared with a newly received version of thesame data sequence (i.e., putatively the same data sequence). Anyreception error will result in a mismatch between the received value andthe corresponding value within the pattern register and therefore, whencompared by XOR gate 513, will result in an error signal being outputfrom the XOR gate 513 to the application logic 515. The applicationlogic 515 may then record the adjusted threshold voltage and clock phaseoffset at which the error occurred to a signal level for a timing offsetwithin a waveform trace.

FIG. 34 illustrates an exemplary waveform trace 527 of a pulse datasequence captured by an embedded scope within the signaling system ofFIG. 33. As shown, a primary pulse 529 arrives at the receiver at symboltime, T₀; a negative reflection 531 of the primary pulse appears atsymbol time T₅ and a positive reflection 533 appears at symbol time T₁₂.Thus, referring to FIG. 33, the application logic 515 of receiver 509may store configuration information in a select logic circuit within theequalizing receiver 510 (or elsewhere within the receive device 509) toenable selection of stored data values having symbol latencies of fiveand twelve symbol times as tap data sources for an equalizing circuit.Alternatively, the application logic 515 may directly output selectsignals to select the desired stored data values as tap data sources.The application logic 515 may also generate tap weights and tap polarityvalues in accordance with the amplitude and polarity of the distortions531 and 533, and store or output the weights and polarity values asnecessary to apply the appropriate tap weights and polarities within theequalizing receiver 510.

FIG. 35 illustrates a method of setting equalization coefficients in asignaling system according to the invention. In the embodiment shown,transmit-side equalization coefficients are set first (541), thenreceive-side equalization coefficients are set (551). The transmit-sidecoefficients are set by transmitting a test signal at 543 (e.g., a pulsesignal, step, etc.), then generating a waveform trace (545) using theembedded scoping techniques described above. The transmit-sideequalization coefficients, including tap data sources, tap weights andtap polarities, are then set at 547 to produce a received waveform tracethat most closely corresponds to the ideal waveform (e.g., pulse, step,etc.) output by the transmitter. The transmit-side equalizationcoefficients may be determined analytically (i.e., by computing thecoefficients based on the waveform trace generated at 545) oriteratively, by repeating operations 543 and 545 for differentcombinations of coefficient settings until a coefficient setting thatprovides a desired waveform is determined.

After the transmit-side equalization coefficients have been set, thereceive-side coefficients are set by transmitting the test signal at 553(i.e., a pulse, step or other signal transmitted with equalizationaccording to the coefficients set at 547), then generating a waveformtrace of the received waveform (555) using the embedded scopingtechniques described above. The receive-side equalization coefficients,including tap data sources, tap weights and tap polarities, are then setat 557 to produce a received waveform that most closely corresponds tothe ideal waveform (i.e., waveform having reduced high-latencydistortion). The receive-side equalization coefficients may bedetermined analytically as described in reference to FIGS. 31-33, oriteratively, by repeating operations 553 and 555 for differentcombinations of coefficient settings until a coefficient setting thatprovides a desired waveform is determined.

Note that selection of tap data sources within the transmitter mayinclude outputting test signals on neighboring signal pathssimultaneously with the test signal transmission at 543 to allowdetermination of which transmit-side equalizer taps, if any, should besourced by cross-talk cancellation data values (i.e., data values beingtransmitted on neighboring signal paths) and the corresponding tapweights.

Reducing Equalization Taps Through Path Length Symmetry.

As discussed above in reference to FIG. 3, the tap select logic 139 andselect circuit 128 enable equalization over a relatively wide range ofsymbol latencies using a small number of equalizer taps. In embodimentsof the invention, the total number of equalizer taps is further reducedthrough symmetry in the electrical distances between signal pathdiscontinuities.

FIG. 36 illustrates a signaling system that employs path length symmetryto reduce the total number of equalization taps needed to compensate forreflection-type ISI. The system includes a pair of circuit boards 571and 573 (e.g., line cards, port cards, memory modules, etc.) havingintegrated circuit (IC) devices 575 and 577 mounted respectivelythereon. IC device 575 includes a transmit circuit coupled to aconnector interface 581 (e.g., a connector or a terminal to be receivedby a connector) via signal path segment 582, and IC device 577 includesa receive circuit coupled to a connector interface 585 via signal pathsegment 586. The connector interfaces 581 and 585 are coupled to oneanother through signal path segment 592 (e.g., backplane trace, cable,etc.) to form an overall signal path between the transmit circuit andreceive circuit.

Because the connector interfaces 581 and 585 tend to have at leastslightly different impedances than the impedance of path segments 582,586 and 592, reflections are produced at connector interfaces as shownby reflection flight paths A_(T), A_(R), C_(T) and C_(R). Morespecifically, the reflection flight path indicated by A_(T) results fromthe primary signal reflecting off connector interface 581, and thereflection reflecting off the output node of the transmit circuit withinIC 575. Thus, the reflection flight time over path A_(T) exceeds theunreflected primary signal flight time by twice the signal propagationtime between the connector interface 581 and the transmit circuit outputnode; i.e., the signal propagation time on path segment 582. Similarly,the reflection flight time over path A_(R) (reflection off receiverinput, then off connector interface 585) exceeds the unreflected primarysignal flight time by twice the signal propagation time between theconnector interface 585 and the receive circuit input; the signalpropagation time on path segment 586. Accordingly, if path segments 582and 586 are designed or calibrated to have equal electrical lengths(i.e., equal signal propagation delays), reflections A_(T) and A_(R)will arrive at the input of the receive circuit of IC device 577 atsubstantially the same time. Consequently, a single equalization taphaving a symbol latency that corresponds to the latent arrival of thecoincident A_(T)/A_(R) reflections may be used to cancel or at leastreduce both reflections. Because reflection flight paths C_(T) and C_(R)are made equal by equalizing the electrical lengths of path segments 582and 586, a single equalization tap that corresponds to the latentarrival of coincident C_(T)/C_(R) reflections may be used to cancel orat least reduce both reflections. Thus, by designing or calibrating pathsegments 582 and 586 to have equal electrical lengths (which pathsegments may optionally include an on-chip path segment between thetransmit circuit output and an IC device 575 output node and/or anon-chip path segment between the receive circuit input and an IC device577 input node), one equalization tap within either the transmit circuitor receive circuit may be used to cancel or reduce a distortion thatwould otherwise require two or more taps.

In one embodiment, the electrical lengths of path segments 582 and 586are made equal (or substantially equal—as achievable through practicablemanufacturing techniques) by design which may include, but is notlimited to: 1) making the physical lengths of path segments 582 and 586substantially equal, whether implemented by printed traces, cables orother types of conductors; 2) including inductive or capacitivestructures (e.g., vias, ferrite materials, narrowed or widened traceregions, or any other impedance-altering structures) statically coupledin series or parallel with path segments 582 and/or 586 to equalizeotherwise different electrical lengths of the path segments; and/or 3)including inductive and/or capacitive structures that may be run-timecoupled (e.g., through pass gates or other electrically or magneticallycontrollable structures) in series or parallel with path segments 582and/or 586 to equalize otherwise different electrical lengths of thepath segments. More generally, any technique for adjusting theelectrical lengths of path segments 582 and 586 to achieve coincidentarrival of two or more signal reflections at the input of an equalizingreceiver may be used without departing from the spirit and scope of thepresent invention.

Regarding run-time coupling of impedance-altering structures to pathsegments 582 and/or 586, such impedance-altering structures may beselectively coupled to path segments 582 and/or 586 through operation ofa configuration circuit (e.g., volatile or non-volatile storage, orfusible or otherwise one-time programmable circuit). For example aconfiguration value that corresponds to the desired electrical length ofa path segment may be programmed into the configuration circuit and usedto control pass gates or other switching elements for switchablycoupling the impedance-altering structures to the path segment. Thedesired setting of the configuration value may be determined, forexample, by using the embedded scoping technique described above inreference to FIGS. 27-29 to determine relative arrival times of signalreflections and therefore propagation time differences between signalreflections.

Although the invention has been described with reference to specificexemplary embodiments thereof, it will be evident that variousmodifications and changes may be made thereto without departing from thebroader spirit and scope of the invention as set forth in the appendedclaims. The specification and drawings are, accordingly, to be regardedin an illustrative rather than a restrictive sense.

1. An apparatus comprising: a scoping circuit to generate a waveformtrace of a test signal received via a signal path, the waveform traceindicating a time interval between receipt of a transition of the testsignal and receipt of a reflection of the transition; a buffer circuitto store a plurality of data values that correspond to data signalsreceived via the signal path; a select circuit coupled to the buffercircuit to select therefrom, according to the time interval indicated bythe waveform trace, a first data value of the plurality of data values;and an equalizing circuit to generate an equalizing signal based, atleast in part, on the first data value.
 2. The apparatus of claim 1wherein the scoping circuit comprises: a signal generator to generate alocal version of the test signal; and a comparator to compare the localversion of the test signal with the test signal received via the signalpath.
 3. The apparatus of claim 2 wherein the scoping circuit furthercomprises a logic circuit to adjust a phase offset of a clock signalused to time reception of the test signal received via the signal path.4. The apparatus of claim 3 wherein the test signal received via thesignal path is a repeating test signal, and wherein the logic circuit isadapted to incrementally adjust the phase offset of the clock signalafter each N repetitions of the test signal, N being an integer greaterthan zero.
 5. The apparatus of claim 4 wherein the each of theincrementally adjusted phase offsets of the clock signal correspond to asample point in the waveform trace.
 6. The apparatus of claim 2 whereinthe scoping circuit further comprises a logic circuit to adjust athreshold voltage, the threshold voltage being used to determine a datavalue represented by the test signal received via the signal path. 7.The apparatus of claim 6 wherein the test signal received via the signalpath is a repeating test signal, and wherein the logic circuit isadapted to incrementally adjust the threshold voltage after each Nrepetitions of the test signal, N being an integer greater than zero. 8.The apparatus of claim 7 wherein logic circuit is adapted to record thethreshold voltage at which the comparator indicates a mismatch betweenthe test signal received via the signal path and the local version ofthe test signal, the threshold voltage at which the comparator indicatesthe mismatch being indicative of the test signal voltage level at aninstant in time.
 9. The apparatus of claim 8 wherein the waveform traceis formed by a sequence of test signal voltage level determinations,each test signal voltage level determination corresponding to athreshold voltage at which the comparator indicates a mismatch betweenthe test signal received via the signal path and the local version ofthe test signal.
 10. The apparatus of claim 1 wherein the scopingcircuit comprises circuitry to generate a select value based on the timeinterval between receipt of the transition of the test signal andreceipt of the reflection of the transition.
 11. The apparatus of claim10 wherein the select circuit is coupled receive the select value fromthe scoping circuit, the select circuit being adapted to select thefirst data value according to the select value.
 12. The apparatus ofclaim 1 further comprising a configuration circuit coupled to thescoping circuit and to the select circuit, the scoping circuit beingadapted to generate a select value based on the time interval and tostore the select value in the configuration circuit, the select circuitbeing adapted to select the first data value based on the select value.13. A method of operation within an integrated circuit device, themethod comprising: generating a waveform trace of a test signal receivedvia a signal path; selecting, based on the waveform trace, a first datavalue of a plurality of data values that correspond to signals receivedvia the signal path; and generating an equalizing signal based on thefirst data value, wherein generating the waveform trace of the testsignal comprises: generating a local version of the test signal; andcomparing the local version of the test signal with the test signalreceived via the signal path, wherein generating the waveform trace ofthe test signal comprises: incrementally adjusting a phase of a clocksignal used to time reception of the test signal to capture samples ofthe test signal at different timing offsets within a window of the testsignal.
 14. The method of claim 13 wherein the test signal is arepeating test signal, and wherein incrementally adjusting the phase ofthe clock signal comprises incrementally adjusting the phase of theclock signal after each N repetitions of the test signal, N being aninteger greater than zero.
 15. The method of claim 14 wherein eachincrementally adjusted phase of the clock signal corresponds to a samplepoint in the waveform trace.
 16. The method of claim 13 whereingenerating the waveform trace of the test signal further comprisesincrementally adjusting a threshold voltage used to determine a datavalue represented by the test signal.
 17. The method of claim 16 whereinthe test signal is a repeating test signal, and wherein incrementallyadjusting the threshold voltage comprises incrementally adjusting thethreshold voltage after each N repetitions of the test signal, N beingan integer greater than zero.
 18. The method of claim 17 furthercomprising recording the threshold voltage at which a comparison of thelocal version of the test signal with the test signal received via thesignal path yields a mismatch, the threshold voltage at which thecomparison yields a mismatch being indicative of a voltage level of thetest signal at an instant in time.
 19. A signaling system comprising: atransmit device including a transmitter, data selector and patterngenerator, the data selector being adapted to select the patterngenerator to provide a sequence of values to be transmitted by thetransmitter during a scoping mode of operation of the transmit device; asignal path coupled to the transmitter of the transmit device; and areceive device including a receiver, receive-side pattern generator,comparator, select circuit, buffer, and equalizer, the receiver beingcoupled to the signal path to receive the sequence of values transmittedby the transmitter, the comparator being coupled to the receiver and thereceive-side pattern generator and adapted to compare the sequence ofvalues received by the receiver with a locally generated sequence ofvalues generated by the receive-side pattern generator, the selectcircuit being adapted to select one of a plurality of storage elementswithin the buffer based on the comparison of the received sequence ofvalues and the locally generated sequence of values, the equalizer beingcoupled to receive a data value from the one of the plurality of storageelements selected by the select circuit and being adapted to generate anequalizing signal based, at least in part, on the data value.
 20. Thesystem of claim 19 wherein the receive device further comprises a logiccircuit to incrementally adjust a phase of a clock signal used to timereception of the sequence of values to achieve a plurality ofincrementally phase-offset samples of each value of the sequence ofvalues.
 21. The system of claim 19 wherein the receive device furthercomprises a logic circuit to incrementally adjust a threshold voltageused to resolve each value of the received sequence of values to one ofa plurality of possible digital values.
 22. A method of operation withina signaling system, the method comprising: receiving a test signal via asignal path; comparing the test signal to a separately generated versionof the test signal to determine a time interval between reception of atransition in the test signal and reception of a reflection of thetransition; selecting one of a plurality of storage elements accordingto the time interval; and generating an equalizing signal based oncontents of the one of the plurality of storage elements.
 23. The methodof claim 22 further comprising transmitting the test signal on thesignal path during a scoping mode of operation within a transmit deviceof the signaling system.
 24. The method of claim 23 wherein transmittingthe test signal on the signal path comprises selecting, within thetransmit device, a pattern generator to provide a sequence of values tobe transmitted as the test signal.
 25. The method of claim 24 whereinthe sequence of values represents a pulse.
 26. The method of claim 24wherein comparing the test signal to the separately generated version ofthe test signal to determine the time interval between reception of thetransition in the test signal and reception of the reflection of thetransition comprises determining a time interval between reception ofthe pulse and reception of a reflection of the pulse.
 27. The method ofclaim 22 wherein comparing the test signal to the separately generatedversion of the test signal to determine the time interval betweenreception of the transition in the test signal and reception of thereflection of the transition comprises: incrementally adjusting a phaseoffset of a clock signal used to capture samples of the test signal; andincrementally adjusting a threshold voltage used to resolve each of thesamples of the test signal to a digital value.
 28. The method of claim27 wherein comparing the test signal to the separately generated versionof the test signal to determine the time interval between reception ofthe transition in the test signal and reception of the reflection of thetransition further comprises generating a waveform trace of the testsignal based on incrementally adjusted threshold voltages at whichsamples of the test signal are resolved to digital values that do notmatch digital values within the separately generated version of the testsignal.
 29. The method of claim 27 wherein comparing the test signal tothe separately generated version of the test signal to determine thetime interval between reception of the transition in the test signal andreception of the reflection of the transition further comprisesdetermining a time interval between a pulse within the waveform traceand a reflection of the pulse.